- •Radio Engineering for Wireless Communication and Sensor Applications
- •Contents
- •Preface
- •Acknowledgments
- •1 Introduction to Radio Waves and Radio Engineering
- •1.1 Radio Waves as a Part of the Electromagnetic Spectrum
- •1.2 What Is Radio Engineering?
- •1.3 Allocation of Radio Frequencies
- •1.4 History of Radio Engineering from Maxwell to the Present
- •2.2 Fields in Media
- •2.3 Boundary Conditions
- •2.4 Helmholtz Equation and Its Plane Wave Solution
- •2.5 Polarization of a Plane Wave
- •2.6 Reflection and Transmission at a Dielectric Interface
- •2.7 Energy and Power
- •3 Transmission Lines and Waveguides
- •3.1 Basic Equations for Transmission Lines and Waveguides
- •3.2 Transverse Electromagnetic Wave Modes
- •3.3 Transverse Electric and Transverse Magnetic Wave Modes
- •3.4 Rectangular Waveguide
- •3.4.1 TE Wave Modes in Rectangular Waveguide
- •3.4.2 TM Wave Modes in Rectangular Waveguide
- •3.5 Circular Waveguide
- •3.6 Optical Fiber
- •3.7 Coaxial Line
- •3.8 Microstrip Line
- •3.9 Wave and Signal Velocities
- •3.10 Transmission Line Model
- •4 Impedance Matching
- •4.1 Reflection from a Mismatched Load
- •4.2 Smith Chart
- •4.3 Matching Methods
- •4.3.1 Matching with Lumped Reactive Elements
- •4.3.4 Resistive Matching
- •5 Microwave Circuit Theory
- •5.1 Impedance and Admittance Matrices
- •5.2 Scattering Matrices
- •5.3 Signal Flow Graph, Transfer Function, and Gain
- •6.1 Power Dividers and Directional Couplers
- •6.1.1 Power Dividers
- •6.1.2 Coupling and Directivity of a Directional Coupler
- •6.1.3 Scattering Matrix of a Directional Coupler
- •6.1.4 Waveguide Directional Couplers
- •6.1.5 Microstrip Directional Couplers
- •6.2 Ferrite Devices
- •6.2.1 Properties of Ferrite Materials
- •6.2.2 Faraday Rotation
- •6.2.3 Isolators
- •6.2.4 Circulators
- •6.3 Other Passive Components and Devices
- •6.3.1 Terminations
- •6.3.2 Attenuators
- •6.3.3 Phase Shifters
- •6.3.4 Connectors and Adapters
- •7 Resonators and Filters
- •7.1 Resonators
- •7.1.1 Resonance Phenomenon
- •7.1.2 Quality Factor
- •7.1.3 Coupled Resonator
- •7.1.4 Transmission Line Section as a Resonator
- •7.1.5 Cavity Resonators
- •7.1.6 Dielectric Resonators
- •7.2 Filters
- •7.2.1 Insertion Loss Method
- •7.2.2 Design of Microwave Filters
- •7.2.3 Practical Microwave Filters
- •8 Circuits Based on Semiconductor Devices
- •8.1 From Electron Tubes to Semiconductor Devices
- •8.2 Important Semiconductor Devices
- •8.2.1 Diodes
- •8.2.2 Transistors
- •8.3 Oscillators
- •8.4 Amplifiers
- •8.4.2 Effect of Nonlinearities and Design of Power Amplifiers
- •8.4.3 Reflection Amplifiers
- •8.5.1 Mixers
- •8.5.2 Frequency Multipliers
- •8.6 Detectors
- •8.7 Monolithic Microwave Circuits
- •9 Antennas
- •9.1 Fundamental Concepts of Antennas
- •9.2 Calculation of Radiation from Antennas
- •9.3 Radiating Current Element
- •9.4 Dipole and Monopole Antennas
- •9.5 Other Wire Antennas
- •9.6 Radiation from Apertures
- •9.7 Horn Antennas
- •9.8 Reflector Antennas
- •9.9 Other Antennas
- •9.10 Antenna Arrays
- •9.11 Matching of Antennas
- •9.12 Link Between Two Antennas
- •10 Propagation of Radio Waves
- •10.1 Environment and Propagation Mechanisms
- •10.2 Tropospheric Attenuation
- •10.4 LOS Path
- •10.5 Reflection from Ground
- •10.6 Multipath Propagation in Cellular Mobile Radio Systems
- •10.7 Propagation Aided by Scattering: Scatter Link
- •10.8 Propagation via Ionosphere
- •11 Radio System
- •11.1 Transmitters and Receivers
- •11.2 Noise
- •11.2.1 Receiver Noise
- •11.2.2 Antenna Noise Temperature
- •11.3 Modulation and Demodulation of Signals
- •11.3.1 Analog Modulation
- •11.3.2 Digital Modulation
- •11.4 Radio Link Budget
- •12 Applications
- •12.1 Broadcasting
- •12.1.1 Broadcasting in Finland
- •12.1.2 Broadcasting Satellites
- •12.2 Radio Link Systems
- •12.2.1 Terrestrial Radio Links
- •12.2.2 Satellite Radio Links
- •12.3 Wireless Local Area Networks
- •12.4 Mobile Communication
- •12.5 Radionavigation
- •12.5.1 Hyperbolic Radionavigation Systems
- •12.5.2 Satellite Navigation Systems
- •12.5.3 Navigation Systems in Aviation
- •12.6 Radar
- •12.6.1 Pulse Radar
- •12.6.2 Doppler Radar
- •12.6.4 Surveillance and Tracking Radars
- •12.7 Remote Sensing
- •12.7.1 Radiometry
- •12.7.2 Total Power Radiometer and Dicke Radiometer
- •12.8 Radio Astronomy
- •12.8.1 Radio Telescopes and Receivers
- •12.8.2 Antenna Temperature of Radio Sources
- •12.8.3 Radio Sources in the Sky
- •12.9 Sensors for Industrial Applications
- •12.9.1 Transmission Sensors
- •12.9.2 Resonators
- •12.9.3 Reflection Sensors
- •12.9.4 Radar Sensors
- •12.9.5 Radiometer Sensors
- •12.9.6 Imaging Sensors
- •12.10 Power Applications
- •12.11 Medical Applications
- •12.11.1 Thermography
- •12.11.2 Diathermy
- •12.11.3 Hyperthermia
- •12.12 Electronic Warfare
- •List of Acronyms
- •About the Authors
- •Index
Transmission Lines and Waveguides |
61 |
1
ln x = x + 1
The solution of this equation is x = ro /r i = 3.591. By substituting this in (3.79), we obtain Z 0 = 76.7V.
3.8 Microstrip Line
A microstrip line consists of a metal strip on one side and a ground plane on the other side of a substrate, as shown in Figure 3.13. The substrate is made of a low-loss dielectric material such as polytetrafluoroethylene (Teflon), aluminum oxide (alumina), or quartz.
A pure TEM wave mode can propagate in a microstrip line only if all fields are in the same medium. Then the solution for the field can be derived from Laplace’s equation. In a case where the nonstatic fields are in two different media, the field has also longitudinal components. At low frequencies, or more precisely when l >> h, the fields are nearly the same as those in a static case, and we call them quasi-TEM. However, the analytical solution
Figure 3.13 The cross section of a microstrip line and the characteristic impedance Z 0 as a function of the ratio of strip width to substrate height w /h for different substrate materials.
62 Radio Engineering for Wireless Communication and Sensor Applications
of the quasi-TEM wave mode is complicated and therefore a practical design of microstrip lines is based on graphs or approximate equations.
The phase constant of a quasi-TEM wave can be expressed as
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me 0 ereff |
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where ereff is the effective relative permittivity. This is obtained by measuring or calculating the capacitance of the line per unit length Ceff and the capacitance of an air-filled but otherwise similar line per unit length C0 :
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The designer often knows the required characteristic impedance Z 0 and the required length of the line in wavelengths l /l. Then the width of
the strip w and the physical length l have to be determined. Z 0 and ereff depend mainly on the width of the strip w and on the height h and the
relative permittivity er of the substrate. The velocity of the wave and the wavelength depend on ereff , as given by (3.83). Sometimes the problem is inverse: The characteristic impedance has to be calculated from the dimensions and permittivity. Approximate design of a microstrip line can be carried out by using Figure 3.13 or the following equations [8]. These equations
are valid when 0.05 ≤ w /h ≤ 20 and er ≤ 16. |
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impedance are |
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Transmission Lines and Waveguides |
63 |
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The width of the strip corresponding to a known Z 0 is obtained for w /h ≤ 2 from
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(3.90)
When w /h ≥ 2,
wh ≈ p2 HB − 1 − ln (2B − 1) + er2−er 1 Fln (B − 1) + 0.39 − 0e.61r GJ (3.91)
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In the preceding equations, it has been assumed that the thickness t of the strip is very small. In practice, the capacitance of the line per unit length will be slightly larger due to the finite thickness t compared to the case t = 0. This will lower the characteristic impedance. Due to the extra capacitance, the width of the strip seems to increase by
Dwe = |
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where D = 2h /t , when w /h ≥ 1/(2p ), and D = 4pw /t, when w /h ≤ 1/(2p). Equation (3.93) is valid for t < h and t < w /2. Equations (3.86), (3.88), (3.89), and (3.91) can be used for a strip having a finite thickness by replacing w by we = w + Dwe .
64 Radio Engineering for Wireless Communication and Sensor Applications
The sources of losses in a microstrip line are:
•Conductor loss in the strip and ground plane;
•Dielectric and conduction losses in the substrate;
•Radiation loss;
•Surface wave loss.
Assuming a constant current distribution over the strip width, the
attenuation constant due to metal loss is |
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The accuracy of this equation is best for a wide strip. In practice, the value of the surface resistance R s is larger than the theoretical one. For example, the surface roughness of the substrate increases R s . The thickness of the conductors should be at least four times the skin depth ds = √2/(vmsm ). A metallization thinner than twice the skin depth would yield excessive attenuation.
Dielectric loss is usually much lower than conductor loss. The attenua-
tion constant due to dielectric loss is |
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ad = p |
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where l0 is the wavelength in free vacuum.
Discontinuities of the microstrip line produce radiation to free space. For a given line, radiation loss increases rapidly as the frequency increases. To avoid leakage and interference, microstrip circuits are usually shielded within a metal case. In microstrip antennas, leaking radiation is harnessed into use.
Surface waves are waves that are trapped by total reflection within the substrate. They may produce unwanted radiation from the edges of the substrate and spurious coupling between circuit elements.
Example 3.5
Find the width of the strip for a 50-V microstrip line. The substrate has a thickness of h = 0.254 mm and a relative permittivity of er = 9.7. The