диафрагмированные волноводные фильтры / 6276a7f1-635a-4317-9366-124168625a32

Abstract—In this paper, a high gain substrate integrated waveguide (S IW) filtering patch antenna with low H-plane cross-polarization (cross-pol) and controllable radiation nulls is proposed. The antenna is composed of a radiation patch integrated with three pairs of shorting pins and a single layer SIW cavity with two trans verse slots. The shorting pins loaded on the patch can improve the gain and reduce the H-plane cross-pol of the antenna. The long slot on the middle layer can not only split the TE110-mode resonator into two half-mode resonators but also generate magnetic coupling between half-mode resonators and the patch. Another short slot achieves mixed electric and magnetic coupling between the left half-mode resonator and the patch that introducing controllable radiation nulls to improve the selectivity. By adjusting the parameters of the short slot, the positions of two radiation nulls can be easily controlled. Finally, one prototype filtering antenna is fabricated and measured for demonstration. Experimental results show that the proposed SIW filtering antenna achieves a high gain of 9 dBi with only one radiator and has the merit of high frequency selectivity as well as low H-plane cross-pol of lower than -43 dB.

Index Terms—Filtering antenna, substrate integrated waveguide (S IW), high gain, radiation null, cross-polarization.

I. INT RODUCT ION

With the rapid development of modern wireless communicat ion technology, the RF front-end system is required to be compact, low cost, high integration and mu lti-functional. So, the antenna with both radiating property and frequency-selective function has become a hot topic. In the past few years, a variety of filtering antennas have been designed and analyzed based on microstrip structures [1]-[7], which have the advantages of simple design, lo w cost and compact structure. For example, by using two microstrip square open-loop resonators, a coupled line, and an L-shaped antenna, a compact filter-antenna can be designed [1]. Multip le radiators can be used to increase the gain and introduce new coupling paths or novel coupling methods [2]-[6]. For instance, a filtering antenna based on metasurface introduces coupling between inner and outer patch cells in a p lane for both high gain and filtering response [6]. But the size of these antennas will increase exponentially. In [7], a novel co mpact high gain differ-

Manuscript received April 26, 2020; revised July 20, 2020. This work is supported by the National Natural Science Foundation of China (NSFC) under grant number 61771360, U19A2055, the Key Industry Chain Project of Shaanxi Province No.2018ZDCXL-GY-08-03-01, the Key Laboratory Foundation No. 6142216180104, the Fundamental Research Funds for the Central Universities and the Innovation Fund of Xidian University. (Corresponding author: Bian Wu)

C. Fan, B. Wu, Y.L. Wang, H.-Y. Xie and T. Su are with the National Key Laboratory of Antennas and Microwave Technology, Shaanxi Joint key Laboratory of Graphene, Xidian University, Xi’an 710071, China (e-mail: bwu@mail.xidian.edu.cn).

entail fed dual-polarized filtering patch antenna is proposed by loading defected ground structures, a cross slot, and shorting pins without extra circuits . Recently, d ielectric resonators are used to design the filtering antenna to avoid the loss caused by the low quality factor of the microstrip structure [8]-[11]. So that the gain of the antenna can be increased effect ively. But the complex modes of the dielectric resonator will increase the design complexity and radiation spurious . Moreover, it also has a high cost. Substrate integrated waveguide (SIW) technology has the merits of low cost and high quality factor, which has advantages to design high-performance filtering antennas [12]-[17]. A mixed frequency-domain/time -domain synthesis technique to integrate SIW filter with a slot antenna and achieve near-lossless transition is first presented in [12]. Based on this, a series of SIW filtering slot antennas have been designed [13]-[16]. For examp le, a half-mode SIW filtering antenna with lo w H-plane cross-pol and good suppression is proposed in [14]. A co-designed SIW filtering antenna with a controllable radiat ion null is proposed, which highly imp roves the selectivity of the filtering antenna [15]. In [16], a third-order SIW-integrated filter/antenna is proposed by putting a slot in the middle of the SIW cavity to realize two resonant modes. Moreover, the resonant frequencies of these two modes can be independently controlled, wh ich can be used to arbitrarily place a null in the antenna gain. But in the present designs, the gain of the filtering antennas barely reaches a high level, the selectivity and cross-pol also can be further improved.

In this paper, a h igh gain SIW filtering patch antenna with low H-plane cross-pol and two controllable radiation nulls is proposed. By introducing mixed electric and magnetic coupling between SIW half-mode resonator and the upper patch, two controllable radiation nulls can be introduced. Meanwhile, an equivalent circuit model is employed to explain the coupling scheme, which agrees well with the fu ll-wave simulat ion model. Moreover, three pairs of short pins are loaded on the patch to improve the gain and reduce H-plane cross-pol level. Finally, a prototype filtering antenna working at 3.5 GHz is designed, fabricated and measured, which ach ieves a peak gain of 9.06 dBi and a low H-p lane cross-pol below -43 d B. Moreover, two radiation nulls are generated at 3.1and 3.9 GHz, lead ing to high selectivity at both lower and upper band edges, with rectangle coefficient given by 1.55 (K = f−35 dB/ f−3 dB).

II.ANT ENNA CONFIGURAT ION AND COUPLING SCHEME

A. Antenna Configuration

The configuration of the proposed filtering antenna is shown in Fig. 1(a), wh ich composed of a coplanar waveguide (CPW) feedline, a single SIW cavity and a patch antenna loaded by three pairs of shorting pins. The SIW cavity is fabricated on the bottom substrate with a permittivity of 2.2 and a thickness of 2 mm. Two transverse slots are etched on the top surface of the

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(a)

Fig. 1.Configuration of the proposed SIW filtering antenna. (a) Exploded view and top view. (b) The bottom surfaces of the SIW cavity. (c) The top surface of the SIW cavity.

SIW cavity to introduce the mixed coupling between the SIW and the patch radiator. The patch has a width of W2, a length of L2, wh ich is fabricated on the F4b substrate with a permittiv ity of 2.2 and a thickness of 5 mm. The top surface of the SIW cavity acts as the ground of the patch radiator. l1 and W1 are the length and width of the patch, d is the diameter of the shorting pins loaded on the patch and lz is the distance between two groups of three shorting pins. Fig. 1(b) shows the bottom surface of the SIW cavity. l3 and W3 are the length and width of the SIW cav ity, respectively. A CPW feed line with a fo lded slot is used to feed the cavity. As illustrate in Fig. 1(c), a long slot and a short slot are located on the top surface of the SIW cavity, with a distance of l6 and l7 from the left edge of the SIW cavity, respectively.

B. Coupling Scheme and Controllable Radiation Nulls

The proposed filtering antenna has three modes , including

two half TE110 modes of the SIW cavity and a TM010 mode of the patch antenna. The long slot with a length of l5 and a width

of W5 can split the TE110 mode SIW resonator into two half-mode resonators, as depicted in Fig. 2. As shown in Fig.3

(b), the resonance frequency of TM010 mode is controlled by l1. The resonance frequency of TM010 mode increases when l1 decreases. The resonance frequency of the left/right half-mode is controlled by l6. As can be seen from Fig.3 (c), when l6 increases, the resonance frequency of the left half-mode decreases but the resonance frequency of the right half-mode increases. Since the electric field of both half-mode resonators reaches the maximu m value at the long slot, so there is an electric coupling between two half modes . When the patch resonates, the magnetic field intensity will reach the maximu m value at the long slot due to the sinusoidal field distribution of TM010 mode. Therefo re, the long slot can introduce a magnetic coupling between the patch and the two half-mode resonators . To provide an insight into the process, a circuit model is presented in Fig. 3(a) to explain the coupling scheme of the filtering antenna with only one long slot. Fig. 3(b) and (c) show

2

(a) (b)

Fig. 2.T he (a) left half-mode and (b) right half-mode of the SIW cavity

Fig.3. (a) Circuit model of filtering antenna with only one long slot. (b) The resonance frequency of TM010 mode with different l1. (c) The resonance frequency of the left/right half-mode of the SIW cavity with different l6. (d) The frequency response of the circuit model. (e) Full-wave simulation results of filtering antenna.

the frequency response of the circuit model and the full-wave simu lation of the filtering antenna with only one long slot. They are in very good agreement with each other, wh ich confirming the validity and accuracy of our circuit model.

As we known, the electric field intensity of TM010 mode will reach the maximu m at the edge of the patch. Therefo re, an electric coupling can be introduced by introducing a slot near the patch edge. As shown in Fig. 1(c), a short slot with the length of l4 and the width of W4 is located on the top surface of the SIW cavity. In this way, the coupling between the left half-mode and the patch will change fro m a magnetic coupling to a mixed electric and magnetic coupling, wh ich can introduce a radiat ion null on each side of the passband. We further modify the circuit model to exp lain the coupling scheme of the filtering antenna with t wo slots, as depicted in Fig.4 (a). The frequency response of the circuit model and the full-wave simulat ion

results of filtering antenna with both slots are given in Fig.4 (b) and (c). The good agreement verifies the validity of the coupling scheme.

According to the resonant theory [18], for the mixed electric and magnetic coupling in Fig.4 (a), when f<f0 (f0 is the center frequency of the filtering antenna), it can be equivalent to an inductance coupling (magnetic coupling), so a radiat ion null (RN1) can be introduced on the left side of the passband. When f>f0, it can be equivalent to a capacitance coupling (electric coupling), so another radiation null (RN2) can be introduced on the right side of the passband. The position of these two radiation nulls can be controlled by changing the intensity of electric coupling or magnetic coupling. Both the short slot and the long slot can control the mixed electric and magnetic coupling, but the parameters of the long slot not only influences the mixed coupling between the half-mode resonator and the patch but also affects the coupling between the two half-mode resonators. So it is better to only change the parameters of the short slot to control these two radiation nulls. Moreover, the simu lated antenna efficiency is also shown in Fig. 4(c). It can be seen that the total efficiency is greater than 90% within passband, which showing good radiating in the passband.

The influence of parameters l7 and W4 on these two radiation nulls is given in Fig.5 (a) and (b), respectively. Fro m wh ich we can conclude that when the nulls are controlled, the antenna is well matched. With the increase of l7, the position of the short

slot will be far away fro m the strongest electric field in the patch center, leading to the enhancement of magnetic coupling. So, RN1 shifts closer to the passband but RN2 remains almost unchanged. Similarly, with the decrease of W4, the electric coupling increases and RN2 shifts closer to the passband while RN1 has little changed.

III.DESIGN AND ANALYSIS OF THE PAT CH ANT ENNA

A. Effect of the shorting pins loaded on the patch

Although the gain of a patch antenna is higher than that of a slot antenna, there is a further requirement for imp rovement . Generally speaking, the length of the rectangular patch antenna is usually related to the half guide-wave wavelength of TM010 mode. So, the gain of the conventional patch antenna is limited to a low level because of their limited effective radiant size at the resonance frequency. Therefore, in the design of the filtering antenna, increasing the number of patches or loading some parasitic patches are effective to increase the gain . However, the co mplex coupling path between these patches will greatly increase the design complexity of filter ing antenna. In this section, three pairs of shorting pins are loaded in the proper positions of the patch to introduce additional inductance.

Fig. 6. T he directivities of the patch antennas without and with shorting pins.

(c)

Fig.7. (a) Surface current distribution without shorting pins, (b) Surface current distribution with shorting pins. (c) Simulated H-plane cross-pol of filtering antennas with and without shorting pins.

The inductance value of the shorting pins can be obtained as:

where h is the thickness of the substrate, 0 is the permeability

in free-space and d is the diameter of the shorting pins, respectively [19]. Due to the shunt inductance effect of these shorting pins, the resonant frequency of the TM010 mode in the patch will increase, and the electrical size of the patch is effectively enlarged and the radiation directivity is hence improved. The directivit ies of the patch loaded without and with shorting pins are depicted in Fig. 6, the directivity can be increased fro m 7.9 d Bi to 9.4 d Bi by loading three pairs of shorting pins. In addition, the loading of shorting pins can also suppress the H-plane cross-pol of the proposed filtering antenna. As described in section II, two coupling slots are located on the top surface of the SIW cavity to introduce a mixed electric and magnetic coupling. As shown in Fig. 7(a), the asymmetry of the slots affects the surface current distribution of the patch at the resonance frequency, which results in an increment of H-p lane cross-pol. However, as we can see from Fig. 7(b ), the existence of shorting pins introduces perturbation to the surface current, making it more symmetrical on the H-plane and thus reducing the cross -pol. Fig. 7(c) depicts the simulated H-plane cross-pol of the filtering antenna with and without shorting pins , which verifies the above analysis.

B. Extraction of the external quality factor of the patch

The filtering antenna is designed by replacing the last-stage resonator of the filter with the patch radiator, so the patch needs to have the same external quality factor (QEXT) as the preceding resonators. Therefore, the extraction of QEXT is a key issue in the replacement of the last-stage resonator in the bandpass filter by a patch radiator. In [2], the QEXT of a slot antenna is extracted for the first time, wh ich has great significance to the design of the proposed filtering antenna. QEXT of a radiator can be expressed by

where QL is loaded quality factor and QU is unloaded quality factor. We assume the radiator to be lossless (QU =∞), QEXT will be approximately equivalent to QL. Fig. 8 (a) shows the simu lation model for QEXT ext raction, the reflection coefficient of the patch can be obtained under the lossless condition. S11min

is the min imu m reflect ion coefficient occurring at the resonant frequency f0, as illustrated in Fig. 8(b). f1 and f2 correspond to

the two frequencies with S11 S11 , where S11 can be defined as:

The coupling coefficient k between the waveguide port and the patch can be found using:

Then, the QEXT of the patch can be obtained by

Frequency(GHz)

(c)

Fig.8. (a) The modelof QEXT extraction. (b) Reflection coefficientsof the model used for QEXT extraction. (c) QEXT of the patch with different W1 and h.

Fig. 8 (c) plots the extracted QEXT of the proposed patch versus its height h with d ifferent width W1. We can conclude that the extracted QEXT of the patch decreases with both the width of the patch W1 and the height of the substrate.

IV. FABRICAT ION AND MEASUREMENT

For validation, a prototype filtering antenna is fabricated. Fig. 9 (a) shows the two sides of the middle layer, two slots are located on this layer for coupling. Fig. 9 (b) shows the bottom and the top layer of the antenna, which are used to feed the SIW resonator and radiate, respectively. As depicts in Fig. 9 (c), the proposed filtering antenna can be assembled by co mbin ing the three layers with a few plastic screws, which do not influence the radiation performance of the filtering antenna.

(c)

Fig.9. Photographs of the fabricated filtering antenna. (a) The middle layer. (b) The bottom and the top layer. (c) The overall structure. (The detailed dimensions of antenna are l1 = 32, W 1= 39, l2 = 58.3, W2= 59, l3 = 42.5, W 3= 43.5, l4 = 1.5, W 4=16.5, l5 = 2.5, W 5=34, l6 = 17.45, l7 = 5.25, lf1 =2, W f1=4.8, lf2 = 15, Wf2= 6.2, lz = 18, d = 2.8. (unit: mm))

that the radiation patterns stable in the passband. The simulated cross-pol in E/ H plane is lower than 25/ 50 d B and the measured cross-pol in E/H plane is lower than20/43 dB. Moreover, the suppression level is more than 26 dB.

In Table I, the performances of the proposed filtering antenna and some previous works are co mpared. This work possesses some advantages in the gain performance, frequency selectivity, electrical size and H-plane cross-pol. Moreover, the proposed filtering antenna introduces two controllable radiat ion nulls by adopting mixed electric and magnetic coupling, wh ich is unprecedented in the design of the filtering antenna.

T ABLE I

COMP ARISON TO PRIOR FILTERING ANTENNA

* Rectangle coefficient of gain: K = f−35 dB/ f−3 dB; CRNs= Controllable radiation nulls;

MPA=Microstrip patch antenna; SA=Slot antenna; DRA= Dielectric resonator antenna;

V. CONCLUSION

In this paper, a high gain SIW filtering antenna with low H-plane cross-pol and two controllable rad iation nulls is proposed. Based on the field analysis of the SIW cavity and the patch radiator, the mixed electric and magnetic coupling is successfully introduced by two slots with d ifferent sizes located on the appropriate positions, which imp rove the selectivity of the antenna effectively. Meanwhile, three pairs of shorting pins are loaded on the patch to improve the gain and reduce the H-plane cross-pol. Measurements indicate that the proposed filtering antenna can achieve a peak gain of 9.06 dBi with only one radiator and also a s maller rectangle coefficient showing higher frequency selectivity. Moreover, the H-plane cross-pol of the antenna is below -43 d B, which is much lower than the previous reported filtering antennas. The proposed structure and the mixed coupling scheme can be potentially extended to design various filtering antennas, which may find applicat ions in the modern RF front-end system.

REFERENCES

[1]W. J. Wu, Y. Z. Yin, S. L. Zuo, Z. Y. Zhang, and J. J. Xie, “A new compact filter-antenna for modern wireless communication systems,”

IEEE Antennas Wireless Propag. Lett., vol. 10. pp. 1131–1134, Oct.2011.

[2]J.-F. Qian, F.-C. Chen, Q.-X. Chu, Q. Xue, and M. J. Lancaster, “A Novel Electric and Magnetic Gap-Coupled Broadband Patch Antenna

With Improved Selectivity and Its Application in MIMO System,” IEEE Trans. Antennas Propag., vol. 66, no. 10, pp. 5625–5629, Oct. 2018.