диафрагмированные волноводные фильтры / 54959ed5-e3b8-4228-a759-47baacf6b045

Abstract— Focal-plane reflector feeds with shared aperture are a promising solution for the increasing number of beams required in the Ka-band satellite services. Stringent manufacturing tolerances and coupling between feeds tend to penalize performance, which is aggravated for a dual band. This paper proposes a new design of a shared aperture antenna intended for multispot single-feed-per beam satellite applications that allows recovering single-feed performance, minimizing the usual impact of adjacent feeds on the field distribution and ultimately on the antenna directivity and gain. We present a new focal-plane dualband Fabry–Pérot cavity antenna (FPCA) design, which allows more reliable fabrication, closer agreement with simulations, and larger bandwidth than the reported solutions. It uses only one double-sided printed frequency-selective surface (FSS) to form the dual-band FPCA, and it is fed through a novel arrangement of double-layer slots in the FPCA-printed ground plane. This favors impedance matching and higher isolation between adjacent feeds. The effectiveness of the solution is experimentally demonstrated for a single-feed Ka-band (20 and 30 GHz) prototype. The generalization for a complete multifeed configuration with filters is shown by full-wave simulations. The decoupling slots on the FPCA ground plane enable restoring single-feed performance in the multifeed system, over the bandwidth of interest.

Index Terms— Closed square loop (CSL), decoupling slots, dual band, Fabry–Pérot cavity antenna (FPCA), feed slots, focal-plane array (FPA), frequency-selective surface (FSS), Ka-band, multiple beams, reflection coefficient.

I. INTRODUCTION

THE increasing bandwidth demand for consumer market satellite broadband communication services is driving large interest and investment on high-throughput Ka-band satellite technology. Most of the current Ka-band satellites implement multiple spot-beam technology to reuse frequency band and/or polarization across the desired coverage area. High-throughput Ka-satellites are pushing for even higher

Manuscript received September 24, 2017; revised December 18, 2018; accepted March 7, 2019. Date of publication April 16, 2019; date of current version July 3, 2019. This work was supported in part by the Fundação para a Ciência e Tecnologia under Project PTDC/EEI-TEL/0805/2012, Project PTDC/EEI-TEL/30323/2017, and Project UID/EEA/50008/2019.

(Corresponding author: Eduardo B. Lima.)

E. B. Lima and C. A. Fernandes are with the Instituto de Telecomunicações, Instituto Superior Técnico, Technical University of Lisbon, 1049-001 Lisbon, Portugal (e-mail: eduardo.lima@lx.it.pt; carlos.fernandes@lx.it.pt).

J. R. Costa is with the Departamento de Ciências e Tecnologias da Informação, Instituto Universitário de Lisboa (ISCTE-IUL), 1649-026 Lisbon, Portugal, and also with the Instituto de Telecomunicações, Instituto Superior Técnico, Technical University of Lisbon, 1049-001 Lisbon, Portugal (e-mail: jorge.costa@lx.it.pt).

Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TAP.2019.2911365

Fig. 1. Satellite simultaneous multispot beam coverage for transmission/reception. (a) Traditional single-feed-per beam. (b) Shared aperture.

capacities and larger coverage areas. Among several means of increasing capacity, the increase in the antenna aperture leads to narrow beam widths, consequently increasing coverage density and the frequency reuse factor [1].

The need for a large number of simultaneous beams, coping with bidirectional communications, low complexity, and high efficiency, ruled out phased-array approaches that are more suited for radio astronomy [2] or for the mobile segment of satellite communications [3].

The common satellite antenna configuration uses a single- feed-per beam at the focal plane, usually horns. However, there is a physical limitation on the number of horns that can be accommodated at the focal plane, since the feed aperture size is constrained both by spillover requirements and by the −3 dB roll-off condition between adjacent beams, see [Fig. 1(a)]. The current/common approach is to significantly increase the number of reflector’s systems to enable full spatial coverage. In this paper, a different solution is adopted, which allows reducing the number of reflectors without compromising performance, concentrating all feeds on a single reflector. This alternative approach is to produce overlapped virtual apertures originating from smaller and closer feeds, see Fig. 1(b). Note that, the coverage must be kept at all times for both transmission and reception, with all beams simultaneously active.

Dielectric lenses might be a solution for sharing the same aperture by multiple miniature feeds integrated at the lens base. Integrated lens antennas have been proposed for satellite-based communication systems [4]–[6] and, namely, for reflector feed [7]. Nevertheless, it represents a bulky solution at the Ka-band.

Much more compact focusing elements have been proposed in the literature, which are compatible with the concept of shared apertures. Solutions based on periodic structures have

been extensively studied to enhance directivity, but mostly single band [8], [9]. Fabry–Pérot cavity antennas (FPCAs) have been adopted for high-directivity designs [10]–[12]. A single-band, single-feed solution is presented in [13], and many single-band multifeed solutions have been proposed in [14]–[17]. Recent developments in antenna arrays with a shared aperture present promising results for scanning applications, for both dual-band operation and triband operation [18], [19]. However, these solutions opt to increase the gap between primary feeds to reduce mutual coupling, leading to sparse arrays, which is not suited for multibeam applications with the stringent beam directivity and separation.

Mutual coupling on heavily populated focal-plane arrays (FPAs) is a major drawback with direct impact on efficiency [20], [22]. This limitation for closely packed arrays is accentuated on a shared aperture solution. When evaluating FPCAs for multibeam applications, not only efficiency is degraded by coupling between feeds but directivity also can be severely affected by the presence of the neighboring feeds. Being a cavity, the field disturbance at the ground plane has a significant impact on the antenna performance and mainly on the beams’ directivity.

Promising single-feed and multifeed FPCA configurations are presented in [23] and [24], respectively. The reported agreement between the measured and predicted gain versus frequency is not remarkable and the results indicate moderately higher losses than desirable, for both single and multifeed configurations.

This paper presents a high-directivity dual-band FPCA that is more robust to fabrication tolerances at the Ka-band, which complies with gain and bandwidth specifications, presents high radiation efficiency and shows reliable performance prediction using standard 3-D electromagnetic (EM) solvers. A multifeed configuration is proposed and analyzed which overcomes the directivity and beam degradation originated by mutual coupling in a shared aperture configuration, recovering the single-feed performance.

This paper is organized as follows. The FPCA design is presented in Section II, and the manufactured prototype and its measurements are shown in Section III. The multifeed configuration is demonstrated in Section IV. The conclusion is drawn in Section V.

II. ANTENNA DESIGN

The FPCA studied in this paper is an open-resonator structure formed by a metallic ground plane and a parallel partially reflective wall (PRW) at height h. In order to be resonant, the cavity must satisfy the following standard condition:

where N is an integer defining the Nth-order cavity mode, and ϕ(λ0) is the phase of the PRW reflection coefficient.

A. Double-Sided FSS on a Single Substrate Layer

Satisfying (1) simultaneously for two different bands with fixed h requires that different electric lengths of the cavity is

Fig. 2. FPCA cut of a unit cell.

Fig. 3. FPCA unit cell dimensions. (a) Top. (b) Bottom.

exactly compensated by ϕ(λ0), the phase of the PRW reflection coefficient at the two bands, see Fig. 2.

Instead of evaluating the phase response at the PRW plane, one can consider ϕ(λ0) together with the phase change due to propagation across the cavity height h, thus considering the PRW structure and the h space as a single block, onward referred as the PRW block (PRWB). Formulation (1) is replaced with

and for simplicity, the left-hand side of (2) is referred to as ϕg(λ0), the PRWB reflection coefficient phase. For any given frequency, in order to have a resonant cavity one must have

A frequency-selective surface (FSS) structure formed with resonant closed square loops (CSLs) is used in this paper as the PRW. The CSL EM behavior has been characterized in [25]. CSL elements are electric resonators with symmetrical geometry, making them suitable for arbitrary polarization.

The adopted unit cell is symmetrical in both xz and yz planes and presents the same period, p, along x and y, see Fig. 3. Unit cell parameters for top (i = 1) and bottom (i = 2) layers are shown in Fig. 3. Parameter ai refers to the CSL outer dimensions, while bi represents the inner dimensions.

The dual-band multibeam array antenna is designed for Ka subbands 19.5–20 GHz and 29.5–30 GHz. Considering the higher frequency wavelength, the cavity height h was set to 6 mm. The substrate is Rogers RT5880 (εr = 2.2 and tanδ = 0.0009), with 1.575 mm (0.062”) thickness.

The reflection coefficient magnitude target is chosen to be between −0.5 and −2 dB to ensure a good antenna impedance matching and to achieve 18 dBi beam directivity at 19.75 GHz and 20 dBi at 29.75 GHz. After parameter optimization, the resulting unit cell dimensions are a1 = 3.3 mm, a2 = 2 mm, b1 = 2.64 mm, b2 = 1.5 mm, and period p is 4.25 mm.

The magnitude and phase of the PRWB reflection coefficient are shown in Fig. 4. Down-link and up-link bands are marked for reference.

Fig. 6. Geometry of the slots etched at the ground plane for the two bands (identical slots on both sides of the dielectric substrate).

B. Fabry–Pérot Cavity Antenna

The double-sided FSS is composed of 19 × 19 cells, to avoid excessive spillover of the coupling slots radiation while ensuring at the same time efficient FSS aperture illumination. The FSS useful size is 80.75 mm × 80.75 mm. A pair of half-wavelength slots is opened at the ground plane for each band, lying parallel to the horn’s H-plane, as in [23]. The ground plane and its coupling slots lie directly at the 10.7 mm × 13 mm aperture of a rectangular horn, fed with linear polarization by a 10.7 mm × 4.3 mm K -band WR-42 waveguide. Fig. 5 shows the E-plane cut of the feeding structure that comprises the horn and the support structure, to which the ground plane with slots is attached. The FSS layer (not shown in Fig. 5) is placed above at height h.

in thickness is used as the ground plane instead of a thin metallic plate. Simulations and measurements have shown that this approach favors wider S11 bandwidth with better impedance matching than obtained with common metallic plate solutions [23]. The FSS layer strongly influences the feed return loss, so the slots must be adjusted considering the entire structure. The obtained slot dimensions are (see Fig. 6): for 20 GHz frequency band, Sl1 = 5.85 mm of slot length, Sw1 = 0.5 mm of slot width, and Sd1 = 11 mm of distance between both slots; for 30 GHz frequency band, Sl2 = 3.8 mm of slot length, Sw2 = 0.75 mm of slot width, and Sd2 = 7.8 mm

Fig. 8. Reflection coefficient of the FPCA. Vertical shaded bars represent Tx and Rx bands.

of distance. Slot dimensions are slightly smaller than if they were opened on a metal plate.

III. MANUFACTURED PROTOTYPE

AND MEASUREMENTS

The fabricated structure is shown in Fig. 7. The FPCA performance is evaluated in terms of impedance matching, realized gain, and phase center. Fig. 8 shows the reflection coefficient versus frequency measured at the waveguide port, compared to simulations; the agreement is quite good. The curves show a comfortably large impedance bandwidth at both frequency bands, covering the intended 19.5 to 20 GHz and the 29.5 to 30 GHz bands.

Fig. 9 presents the simulated and measured realized gain curves versus frequency. Agreement between the simulated and measured curves is quite good again. Some ripple appears for higher frequencies due to lower measurement dynamic range. The lower band presents its maximum of 16.2 dBi at 19.8 GHz, with 4.5% bandwidth and at higher band 18.5 dBi

Fig. 9. Realized gain of the FPCA. Vertical shaded bars represent Tx and Rx bands.

Fig. 10. Magnitude of the FPCA radiation pattern at 19.75 GHz (black dashed curve—measured co-pol, blue line—simulated co-pol, and green round dot— measured X-pol). (a) H-plane. (b) E-plane.

Fig. 11. Magnitude of the FPCA radiation pattern at 29.75 GHz (black dashed curve—measured co-pol, blue line—simulated co-pol, and green round dot— measured X-pol). (a) H-plane. (b) E-plane.

Fig. 12. Phase of the FPCA co-pol radiation pattern at 19.75 GHz (black dashed curve—measured and blue line—simulated). (a) H-plane. (b) E-plane.

at 29.5 GHz, with 2.4% bandwidth. The desired bandwidth is covered in both frequency bands.

The simulated and measured return loss are significantly high in [23], reaching −4 dB at high band. The FPCA design presented in [23] and [24] is a narrowband solution at both bands, not covering the required bandwidth, being very sensitive to any frequency deviation. The use of etched slots on a double-sided metalized substrate enables to extend the feed bandwidth at both bands, crucial to accommodate fabrication errors or deviations, see Fig. 8. Bandwidth is larger than 1 GHz at both bands (−10 dB level), allowing small frequency deviations without performance degradation. The proposed ground plane configuration, with slots on a double-sided metalized substrate, enables close multiresonances, unlike the simple metal layer with etched slots, favoring wider matching bandwidth.

The measured H- and E-plane radiation patterns at 19.75 and 29.75 GHz are superimposed on simulations in Figs. 10 and 11, respectively. Good agreement is achieved, and the cross-polarization is always below −20 dB, showing an almost perfect linear polarization.

The measured phase pattern is shown in Figs. 12 and 13. An almost flat phase is achieved in the main lobe, with simulations and measurements presenting the same behavior. The phase center is similar in simulations and measurements, being approximately 60 mm behind the antenna cavity. Although the radiation patterns in Figs. 11–13 are only shown for the center frequencies of each bandwidth, measurements were made also

Fig. 13. Phase of the FPCA co-pol radiation pattern at 29.75 GHz (black dashed curve—measured and blue line—simulated). (a) H-plane. (b) E-plane.

for the bandwidth edges with similar good agreement with simulations. Coincident phase center for both frequency bands is crucial as a reflector dual-band primary feed, in order to maximize the reflector aperture efficiency and optimize scanning performance.

IV. MULTIFEED CONFIGURATION

The shared aperture concept is intended to pack the individual feeds very tightly to increase the beam spot density. However, the mutual influence between the feeds may defeat the objective, causing severe degradation of the individual feeds’ performance. It not only affects the individual feed

Fig. 14. Seven-feed subsets of a larger aperture. The central feed and its neighboring elements with the same polarization are aligned along its

(a) E-Plane and (b) H-plane.

Fig. 15. Dual-band filter. (a) Fully assembled prototype. (b) 20 GHz SRRs. (c) 30 GHz SRRs.

gain, by increased losses due to coupling but also decreases the directivity by reducing the aperture efficiency. To the best of the authors’ knowledge, this has not yet been completely solved in the literature for dual-band operation.

Possible strategies to isolate the feeds’ response rely on the use of polarization diversity and frequency reuse inside the band, resorting to sharp filters to create adjacent subbands. Orthogonally polarized feeds present negligible coupling and consequently small interference on the feed performance. However, feeds with the same polarization, although more distant from each other, still show mutual influence. Even the use of passband filters is not sufficient to provide the needed isolation. The EM field that is blocked by the filters is reflected back into the FSS with arbitrary phase and parallel slots from different feeds couple between each other.

A. Polarization Diversity

For simplicity, only a subset formed by seven feeds of a larger aperture is evaluated assuming that further away feeds have minor relevance for the central feed performance. A common feed arrangement with polarization diversity is shown in Fig. 14. Feeds with the same polarization are placed along the same plane. Two different subsets of the aperture need to be evaluated: the case where the central feed plus the two adjacent feeds with the same polarization are aligned along the E-plane, Fig. 14(a), and the case where the three feeds are aligned along the H-plane, see Fig. 14(b). In a large array, both configurations repeat periodically. Note that, results presented in [24] refer to only one of these cases, thus lacking part of the system performance information.

The fL label designates the low-frequency subband while fH refers to the high-frequency subband. Each configuration shown in Fig. 14 must be evaluated considering both fL and fH subbands’ for the central feed. This actually translates into four independent cases that must be evaluated and optimized individually.

B. Frequency Diversity

Filters can be used to allow the constructive sum of all reflected fields from out-of-band FPCA feeds with the same polarization.

A dual-band miniaturized filter was designed by the authors to work at both 20 and 30GHz subbands [26], in order to maintain the FPCA performance in a multifeed configuration,

Fig. 16. Simulated filter magnitude response. (a) Low band filter.

(b) High band filter.

see Fig. 15. It uses the transmission characteristics of a cutoff rectangular waveguide periodically loaded in the E-plane with split-ring resonators (SRRs). The filter is terminated at both ports with a coaxial cable probe. The dual-band filter was designed to produce 200 MHz wide lower subbands (19.5–19.7 and 29.5–29.7 GHz). The filter for the upper subbands (19.8–20 and 29.8–30 GHz) is just a frequency shifted up version of the designed and fabricated one.

The filter’s full-wave simulated magnitude and phase performance are shown in Figs. 16 and 17, respectively. The insertion loss is around 2 dB inside the filter passband and close to 1 dB outside. The moderately high insertion loss is a direct consequence of the used materials; nevertheless, it still serves as proof-of-concept. Out of the band phase variation has a large slope, which will translate into narrowband performance at each frequency subband, as will be shown ahead.

Each feed’s waveguide of the FPCA structure is terminated by a waveguide-to-coaxial transition (WCT), compatible with the filter termination, see Fig. 18. Thus, a waveguide cavity

Fig. 17. Simulated S22 filter phase response.

Fig. 18. Feed structure cut along the E-plane.

is formed between the horn closed (slotted) aperture and the WCT shorting wall. The probe diameter is = 0.51 mm, the same as the filter probe. Proper dimensioning of the waveguide (WL ) and coaxial cable probe (lL B and lH B ) lengths for both the low and high subbands, as well as the probe’s distance to the back wall, enables obtaining constructive interference of all the reflected fields when they reach each horn’s slotted aperture.

C. Co-Polar Slot Isolation

Despite the use of frequency diversity, the coupling is still significant between slots with the same polarization. The proposed solution to overcome this coupling is to include both E- and H-plane slots, see Fig. 19. The inclusion of extra slots, between feeds, allows increasing the illuminated area of the FPCA, with direct impact on directivity. Note that, these isolating slots are only added at the upper side of the dielectric substrate, keeping a completely closed ground plane at the bottom side. Coupling between the 20 GHz slots is considerably stronger when these are aligned along the E-plane, the configuration of Fig. 14(a). Isolation slots placed along the E-plane allow recovering the 20 GHz band performance, while the 30 GHz higher directivity is reasonably kept by inserting slots along the H-plane.

As an example, considering tight feeds, a separation between feed-array elements d = 20 mm was assumed (1.3λ0 and 2λ0 at 20 and 30 GHz, respectively). SwE , SwH and SlE , SlH represent the isolation slots’ width and length, respectively. The isolation slots are equally separated by SdE = SdH = d, see Fig. 19.

D. Multifeed FPCA Implementation and Evaluation

Simulation of the full structure composed of seven FPCA in combination with all filters was impossible to perform

Fig. 19. CST upper layer model with slots added at both the E- and H-planes. Feeds positioned along (a) E-plane and (b) H-plane.

Fig. 20. CST schematic of the multifeed configuration, including the subbands’ filters ( fL case—port 1) [27].

with available in-house resources and also because different solvers were used for both cases: frequency solver for the filter simulation and transient solver for the multifeed FPCA. Therefore, a combined co-simulation process was adopted. Both filters were replaced with their previously simulated S-parameters matrices and the multifeed FPCA was simulated with the CST schematic, see Fig. 20. Four combinations were evaluated as already explained: with fL as a reference in both the E-plane and H-plane feeds’ distribution and fH in the same circumstances.

All the structure dimensions were optimized considering directivity maximization for both the E- and H-plane configurations and for both low and high subbands. The resulting waveguide length, including waveguide-to-coaxial cable transition, is 30.8 mm. Coaxial cable probe length and distance to back wall are 2.7 and 2.3 mm, respectively. A tuning pin with= 1.06 mm in diameter and 0.3 mm in height was added 3 mm away from the waveguide back wall, to compensate for probe reactance.

Coaxial cable length was defined separately for the low and high subbands, enforcing a different phase shift on the reflected wave for the out of the band filter. The obtained lengths, lL B = 15.9 mm (low subbands) and lH B = 14.9 mm (high subbands), correspond to a compromise between all configurations. At last, the resulting dimensions of the

Fig. 22. Reflection coefficient of the multibeam FPCA with feeds positioned along the E-plane, vertical shaded bars represent Tx and Rx bands.

(a) 20 GHz. (b) 30 GHz.

ground plane orthogonal isolation slots were SwE = SwH = 0.5 mm, SlE = 10 mm, and SlH = 5 mm.

To illustrate the relevance of the extra slots on the upper side of the ground layer, Fig. 21 shows the comparison of the simulated E-field with and without the isolation slots considering all seven feeds. When the slots are included, it is visible that the reflected field inside the neighboring waveguides is stronger and the illuminated FPCA top layer area is larger, favoring higher directivity. The interpretation is that in the second case, Fig. 21(b), the field inside the other waveguides is the outcome of reflections at the top layer, with the proper phase, and not from coupling, as happens when the isolation slots are not included.

Coupling between feeds is kept marginal at the frequency subbands due to the filter response. S11 is kept below −10 dB

Fig. 23. Simulated directivity with feeds positioned along (a) E-plane and

(b) H-plane.

Fig. 24. Simulated directivity for multifeed configurations to evaluate the filter’s and slot’s influence. Feeds placed along (a) E-plane and (b) H-plane.

at both frequency subbands and for both polarizations. Feeds placed along E- and H-plane have similar S11, and therefore, only the E-plane case is shown, see Fig. 22.

The simulated directivity is superimposed in Fig. 23 on the single feed and on the multifeed results without using filter and isolation slots. Two cases are shown: feeds along the E-plane and along the H-plane. In the interval of relevance, marked

in red, directivity is recovered to single-feed levels but the filter large phase slope and the not perfect response of the filter in magnitude, enforce a narrower band, especially close to the center of both 20 and 30 GHz bands. Nevertheless, it is demonstrated that single-feed performance can be obtained in a multifeed configuration with a shared aperture. An improvement between 1 and 2 dB is obtained in comparison with the regular multifeed configurations.

In order to individually evaluate the filters’ and slots’ influence, the simulated directivity for each case is shown in Fig. 24. It is important to note that with the feeds placed along the H-plane, the presence of the slots is neglectable, but when placed along the E-plane the improvement is significant, around 1 dB.

V. CONCLUSION

A new approach is proposed for a dual-band FPCA, intended for shared aperture focal-plane reflector feeding. A prototype was manufactured and directly verified experimentally at the Ka-band. The implementation of feed slots printed in a dielectric with double-sided metallization considerably improves the impedance matching in comparison to commonly used metal crafted slots.

A new strategy was proposed and analyzed by simulation to operate with multiple closely spaced feeds sharing the same FSS. Further to the usual polarization diversity and frequency reuse strategies, the proposed configuration defeats the known limitations due to mutual coupling and successfully allows recovering the single-feed beam directivity within predefined low and high-frequency subbands.

A complete feasibility study for a focal-plane shared multifeed structure was presented. A new FSS was proposed, which is robust to fabrication tolerances at mm-waves. Finally, a solution was presented to closely pack multiple feeds without sacrificing the individual feed performance.

ACKNOWLEDGMENTS

The authors would like thank J. Farinha and C. Brito for the prototype construction. They would also like thank A. Almeida for the prototype measurements.

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Eduardo B. Lima (S’14) received the Licenciado, M.Sc., and Ph.D. degrees in electrical and computer engineering from the Instituto Superior Técnico (IST), Technical University of Lisbon, Lisbon, Portugal, in 2003, 2008, and 2018, respectively.

From 2004 to 2015, he was a Research Engineer with the Instituto de Telecomunicações (IT), Lisbon. Since 2015, he has been as an Antenna Engineer. He has coauthored over eight patent applications and 50 technical papers in international journals and conference proceedings in the area of antennas.

Dr. Lima was a TPC member at EuCAP2015. He received the CST University Publication Award in 2010. He is currently a Technical Reviewer for the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION.

Jorge R. Costa (S’97–M’03–SM’09) was born in Lisbon, Portugal, in 1974. He received the Licenciado and Ph.D. degrees in electrical and computer engineering from the Instituto Superior Técnico (IST), Technical University of Lisbon, Lisbon, in 1997 and 2002, respectively.

He is currently a Researcher with the Instituto de Telecomunicações, Lisbon. He is also an Associate Professor with the Departamento de Ciências e Tecnologias da Informação, Instituto Universitário de Lisboa (ISCTE-IUL), Lisbon. He has coauthored

over four patent applications and 150 contributions to peer-reviewed journals and international conference proceedings. More than 30 of these papers have appeared in IEEE journals. His current research interests include lenses, reconfigurable antennas, MEMS switches, UWB, multi-in multi-out (MIMO), and RFID antennas.

Dr. Costa served as an Associate Editor for the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION from 2010 to 2016 and he was a Guest Editor of the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION Special Issue on Antennas and Propagation at mmand Sub-mm-Waves in 2013. He was the Co-Chair of the Technical Program Committee of the European Conference on Antennas and Propagation (EuCAP 2015) in Lisbon and a General Vice-Chair of EuCAP 2017 in Paris.

Carlos A. Fernandes (S’86–M’89–SM’08) received the Licenciado, M.Sc., and Ph.D. degrees in electrical and computer engineering from the Instituto Superior Técnico (IST), Technical University of Lisbon, Lisbon, Portugal, in 1980, 1985, and 1990, respectively.

He joined IST in 1980, where he is currently a Full Professor with the Department of Electrical and Computer Engineering, focusing on microwaves, radio wave propagation, and antennas. He is currently a Senior Researcher with the Instituto de

Telecomunicações and a member of the Board of Directors. He has coauthored a book and 130 technical papers in peer-reviewed international journals and conference proceedings in the areas of antennas and radio-wave propagation modeling. His current research interests include dielectric antennas for millimeter-wave applications, antennas and propagation modeling for personal communication systems, RFID antennas, artificial dielectrics, and metamaterials.

Dr. Fernandes has been the leader of antenna activities in national and European Projects such as RACE 2067-MBS (Mobile Broadband System), ACTS AC230-SAMBA (System for Advanced Mobile Broadband Applications), and ESA/ESTEC-ILASH (Integrated Lens Antenna Shaping). He was a Guest Editor of the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION

Special Issue on Antennas and Propagation at mmand Sub mm-Waves in 2013.