диафрагмированные волноводные фильтры / 77071c35-eaae-407f-ba2e-3180105164aa

Fig. 1. (a) Configuration of the SIW filtering antenna (dimensions: s = 0.2, p = 4, W = 20, r1 = 2, r2 = 4.5, l = 1.6 unit: mm) and (b) a photograph of the fabricated prototype.

Fig. 2. Variation of |S11| with (a) cavity size W (solid line: W = 20 without pin, dashed line: W = 20 with pin, dashed dot line: W = 20.8 with pin, unit: mm) and

(b) offset p.

tanδ = 0.0002. An annular ring slot of inner and outer radii of r1 and r2, respectively, is etched on the top wall for radiation.

Any two orthogonal degenerate resonant modes like TEmn0

and TEpq0 of a SIW resonator for a given width W and length L of a dual-mode cavity are related as

The modes TEmn0 and TEpq0 are orthogonal to each other and hence intermodal coupling does not exist. Coupling between the

modes is obtained by introducing some sort of geometrical perturbation such as metallic via, corner cut, slot on top plane etc. Here, a vertical corner metallic pin at diagonal offset p from the center is used for the perturbation. The degenerate modes used here are the first higher order modes next to the fundamental mode. This mode has a major advantage over the fundamental mode in terms of less conductor loss due to evenly distributed current density [5].

Perturbation results in mode splitting. Fig. 2(a) shows the variation of |S11| with and without the metal pin. The first dip is

due to the fundamental TE110 mode. The second dip is due to the TE210/TE120 mode, which splits into two modes when the metal

pin is introduced. Since higher order mode is being used, it is observed that the pin with diameter 0.8 mm at p = 4 mm decreases the inductance of the equivalent parallel resonator. Therefore, the cavity size should be readjusted to bring back the

Fig. 3. Time averaged electric field distributions (a) inside the cavity, and (b) on the slot plane.

Fig. 4. Variation of |S11| with feed inset depth l for p = 4 mm, (a) f210 mode and

(b) both the modes.

Fig. 5. (a) Realized gain variation with l for p = 4 mm and for θ=0°, φ=0°, (b) The expanded view for f210 band.

resonant frequency. Here, a cavity with pin and W = 20.8 mm is equivalent to a cavity without pin with W = 20 mm. The offset p controls the coupling between the TE210/TE120 modes and hence the separation between f210 and f120. Fig. 2(b) shows the variation of f120/f210 with p. Maximum separation between the two bands is achieved for p = 3.5 mm.

Fig. 3 shows the electric field distributions at the two resonant frequencies. The electric field inside the cavity is mostly Z-directed as shown in Fig. 3(a). Fig. 3(b) shows the time averaged electric field distributions on slot plane. It shows that

the mode with f210 = 10.47 GHz is due to odd-mode resonance since the corner pin is at null and an electric wall can be placed

along the diagonal of the cavity passing through the pin. Therefore, f210 is not affected by p. Similarly, f120 = 11.87 GHz is due to even-mode resonance since a magnetic wall can be placed along the same diagonal plane. Therefore, f120 is

controllable by p. At the odd-mode frequency f210, ZX-plane is the E-plane as the electric field variation is along X-axis on the

slot plane. Similarly, for the even mode frequency f120, YZ-plane is the E-plane.

Fig. 6. Simulated and measured realized gain patterns at (a) 10.47 GHz on ZX- plane (E-plane) and YZ-plane (H-plane) and (b) 11.87 GHz on YZ-plane (E- plane) and ZX-plane (H-plane) (solid line, circle: E-plane, dashed line,star: H- plane).

External quality factor Qext determines the input matching bandwidth and is controlled by the feed inset depth l. The plot

of |S11| for f210 mode in Fig. 4(a) shows that l can be used to control the 10 dB input impedance matching bandwidth. The

optimum feed offset l is first determined for best impedance matching at the design frequencies. Any deviation from the optimum l value increases the Qext and hence decreases the effective bandwidth of the antenna. Next, keeping the bandwidth fixed, matching can be further improved, if required, by varying s. Fig. 4(b) shows that almost equal matching is obtained for both the modes for l = 1.6 mm and s = 0.2 mm. Measured |S11| is also shown in Fig. 4(b). Claim of bandwidth control is also supported by the plots of realized gains shown in Fig. 5(a). The expanded view for f210 band is shown in Fig. 5(b) for clarity. Pass bands are centered at 10.47 GHz and 11.87 GHz.

III. RESULTS AND DISCUSSION

Simulated and measured radiation patterns of co-pol and cross pol on ZX and YZ-planes are shown in Fig. 6(a) and Fig. 6(b) at 10.47 GHz and 11.87 GHz, respectively. The antenna is theta polarized on XZ-plane at 10.47 GHz and phi polarized at 11.87 GHz. The antenna is thus capable of radiating in dualband with dual-polarization.

IV. CONCLUSION

A dual-band dual-polarized filtering antenna has been designed using the concept of first higher order degenerate mode splitting and a technique is described to control the bandwidths. Therefore, the antenna does not require external bandpass filter, thus, reducing the system cost and volume. The high selectivity of the filtering antenna makes it suitable for several narrow band secure antenna applications.

REFERENCES

[1]Y. Wang and Z. Du, “Dual-polarized dual-band microstrip antenna with similar-shaped radiation pattern,” IEEE Trans. Antennas Propag., vol. 63, no. 12, pp. 5923-5928, Dec. 2015.

[2]H. Lee, Y. Sung, C. M. Wu and T. Itoh, “Dual-band and polarizationflexible cavity antenna based on substrate integrated waveguide,” IEEE Ant. Wireless Propag. Lett., vol. 15, pp. 488-491, 2016.

[3]J. Lin, Z. Qian, W. Cao, S. Shi, Q. Wang and W. Zhong, “A low-profile dual-band dual-mode and dual-polarized antenna based on amc,” IEEE Ant. Wireless Propag. Lett., vol. 16, pp. 2473-2476, 2017.

[4]C.Y Chang and W. C C Hsu, “Novel planar, square-shaped, dielectricwaveguide, single-, and dual-mode filters,” IEEE Trans. Microw. Theory Techn., vol. 50, no. 11, pp. 2527-2536, Nov. 2002.

[5]A. A. Khan and M. K. Mandal, “Compact self-diplexing antenna using dual mode SIW square cavity,” IEEE Ant. Wireless Propag. Lett., vol. 18, no. 2, pp. 343-347, Feb. 2019.