диафрагмированные волноводные фильтры / 766641d7-1454-407a-b51f-26dbe085099e

Figure 8 Measured and simulated results of the filter with triple-mode and circular dual-mode cavities

attractive because of its planar structure, low cost, low insertion loss, and flat pass-band. It is then combined with a circular dualmode cavity to obtain a highly selective filter with better out-of- band rejection. The agreement between simulation and measured results validates the proposed structure. The scheme in this letter is very suitable for high-frequency application.

ACKNOWLEDGMENTS

This work is supported by National Science Foundation under Grant NSFC60621002.

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12.Y.D. Dong, W. Hong, and H.J. Tang, Millimeter-wave dual-mode filter using circular high order mode cavity, Electron Lett, submitted.

© 2009 Wiley Periodicals, Inc.

A MINIATURIZED DUAL-MODE ZEROTH-ORDER RING BANDPASS FILTER WITH ONE LEFT-HANDED UNIT AND A CAPACITIVE STUB

Lin-Sheng Wu, Xi-Lang Zhou, Wen-Yan Yin, and Jia-Xiao Niu

Center for Microwave and RF Technologies, Shanghai Jiao Tong University, Shanghai 200240, People’s Republic of China; Corresponding author: wallish0@hotmail.com

Received 4 July 2008

ABSTRACT: A novel miniaturized dual-mode ring bandpass filter is presented. By using only one resonant-type left-handed unit with complementary split ring resonator to combine with a microstrip line, the ring configuration provides two orthogonal zeroth-order resonance modes and its occupied area can be reduced by 82% when compared with the conventional one. A capacitive stub is also introduced to control the transmission zero frequencies. The spurious suppression characteristics is also improved. The filter performances are demonstrated by our measured results, with good agreement when compared with the simulated ones. © 2009 Wiley Periodicals, Inc. Microwave Opt Technol Lett 51: 603– 607, 2009; Published online in Wiley InterScience (www. interscience.wiley.com). DOI 10.1002/mop.24113

Key words: ring bandpass filter; dual-mode; left-handed; zeroth-order resonance; miniaturization

1. INTRODUCTION

Since the dual-mode ring bandpass filter (BPF) was proposed [1] in 1972, various designs have been realized because of its attractive features [2–11]. For the conventional ring BPF, the circumference is about one guided wavelength at its operating frequency. Therefore, its size may be large when the frequency is low, and its miniaturization is an important consideration for practical applications. More recently, several novel techniques have been developed for reducing the size of the ring, such as the meander loop [3], shunting equivalent capacitance [4 – 6] or low impedance section [7] along the ring circumference, triangular [8] or hexagonal loop [9], etching patterns on the ground plane [5, 10], and applying microstrip-to-coplanar waveguide broadside-coupled section [11], etc. The above-mentioned designs show the best area reduction up to 80% compared with the traditional ring BPF. However, all first-order ring BPFs are still restricted by their operating wavelength, respectively.

To the best of our knowledge, the dual-mode zeroth-order ring configuration was firstly proposed [12] in 2005, which was constructed by composite right/left handed (CRLH) transmission lines implemented with quasi-lumped elements of interdigital capacitors and short-circuit stub inductors, and supported two orthogonal unique zeroth-order modes. A dual-mode zeroth-order ring resonator implemented with lumped-element CRLH transmission lines was also presented and exhibited the elimination of the resonant mode at the second harmonic [13]. Besides, a left-handed (LH)

L-band notch bandstop filter based on the zeroth-order ring resonator was designed with significantly reduced size [14]. But till now, no CRLH ring BPF has been realized yet.

In this article, a dual-mode zeroth-order ring bandpass filter with only one resonant-type left-handed unit is proposed. Further, a capacitive stub is loaded on the symmetry plane of the structure so as to control the transmission zero frequencies. By applying left-handed unit, the filter not only has the first spurious response 2.58 times away from the central frequency but also an area reduction of 82% when compared with the conventional ring filter.

2. DESIGN AND ANALYSIS

2.1. Resonant-Type Left-Handed Unit with Complementary Split Ring Resonator (CSRR)

The structure consisting of complementary split ring resonators and microstrip series gap can produce LH transmission band [15]. Because the LH passband is located near the resonant frequency of CSRR etched on the ground plane, this type of LH structures are called resonant-type left-handed unit. Without any lumped elements and grounded via-holes, this component can be fabricated only through the photo-etching technique and no additional process is needed, and it is of course convenient for manufacturing.

Its configuration and simulated S-parameters are shown in Figures 1 and 2, respectively, where the relative dielectric constant of the substrate is r 2.65, the thickness is h 0.5 mm, and

means the LH unit can provide negative phase-shifting characteristics in its passband.

This attractive feature has been used to miniaturize some microwave circuits. In [16], the researchers replace the 270° branch line of the conventional hybrid ring by a 90° left-handed branch line realized by two CSRR units. Under such circumstances, the occupied area can be reduced by 60% when compared with the

Figure 1 The configuration of resonant-type left-handed unit (black for top metal and grey for CSRR on the bottom ground)

Figure 2 Simulated results of the left-handed unit (obtained from Ansoft HFSS)

conventional one, and this technique can be also introduced into the design of dual-mode ring BPFs.

2.2. Dual-Mode Zeroth-Order Ring Resonator

Based on the theory of Hill’s operator, the resonant frequencies in the ring resonator are determined by the ring structure, which is characterized by potential, q x , and the resonant frequencies in a ring resonator coincide with the square root of eigenvalues of the corresponding Hill’s equation [17]. For conventional ring resonators, the first pair of resonant frequencies are corresponding to the two modes degenerated from the first eigenvalue of the second order. Their electric circumferences are about 2 , whereas the real circumferences are about one guided wavelength. However, when the LH transmission lines are introduced to build zeroth-order ring resonators, the first pair of resonant frequencies is corresponding to the two orthogonal modes of the eigenvalue near zero which is usually corresponding to direct current in conventional ring resonators.

In [12], the zeroth-order ring structure is composed of eight uniform quasi-lumped CRLH units which are radial symmetric to the center. The two zero phase origins between bandgap and passbands are utilized to provide zero phase-shifting characteristics, so two valid resonance modes called voltage mode and current mode arise along with their corresponding frequencies.

Dissimilarly, to reduce the size of component, only one LH unit are used in our design to build the zeroth-order ring resonator together with a microstrip transmission line, and its electric length satisfies:

where lRH is the real length of the microstrip line, RH is the guided-wave velocity, and the geometry is shown in Figure 3.

Obviously, the resonance will arise near the resonant frequency of the LH unit. Because the geometry is symmetric and the introduction of single LH unit causes nonuniformity, the zerothorder resonance will degenerate into two orthogonal modes, odd mode with symmetry electric wall and even mode with symmetry magnetic wall. Because the nonlinearity of the LH unit is greatest in the ring configuration, the energy will concentrate in it, which affects the distribution of the electromagnetic fields responding to the even mode. Thus, the odd mode dominates in the CRLH ring resonator while the even mode is relatively minor.

Figure 3 Geometry of the CRLH ring resonator

2.3. Dual-Mode Zeroth-Order CRLH Ring Bandpass Filter

Because the CRLH ring resonator is much smaller than the conventional ones, it cannot provide enough excitation by coplanar coupled microstrip lines. Thus, the resonator is fed by a pair of 50 coplanar waveguide (CPW) feeding-lines on the bottom plane, while the ring resonator is located on the top plane, as shown in Figure 4.

The condition to generate transmission zeros of a dual-mode ring BPF is given by [18]:

where y21,LH and y21,RH represent the transfer admittances of the leftand right-handed parts of the ring structure, respectively. In fact, to

select the location of I/O terminals properly can provide two

transmission zeros for the dual-mode ring BPF. Here, the parameter l10 can be used for this purpose, while it also affects the transmission poles in the passband.

2.4. Capacitive Stub Loaded

In [18], a novel method of inducing a stub perturbation at the end of symmetry plane is proposed to control the transmission zero frequencies of a dual-mode ring BPF, keeping the bandwidth constant. Similar to [18], an open-ended capacitive stub is loaded on the symmetry plane in our design for controlling the transmission zeros more freely, whose equivalent electric length is always less than quarter a guided wavelength, shown in Figure 5.

Obviously, the resonant frequency of odd mode is rarely affected by the capacitive stub loaded, whereas the resonant frequency of even mode should be decreased because the stub increases its corresponding effective electric length. Because y21,RH is affected by the capacitance loaded, the two transmission zero frequencies will also change. Tuning the location of I/O terminals simultaneously, the overall characteristics will be optimized.

3. EXPERIMENTAL RESULTS AND DISCUSSION

3.1. Dual-Mode Zeroth-Order CRLH Ring Bandpass Filter with Only One LH Unit

The proposed dual-mode zeroth-order CRLH ring bandpass filter with only one LH unit was fabricated, shown in Figure 6(a), with

contactors on the I/O terminals. The S-parameters are plotted in Figure 6(b), with the simulated ones carried out using Ansoft HFSS, a full-wave simulator, and good agreement achieved.

The central frequency of the fabricated BPF is 1.94 GHz, where S21 is 1.2 dB, and the 3-dB bandwidth is about 180 MHz. There are two transmission zeros at 2.18 GHz and 2.67 GHz, respectively, which can improve the rejection performance in the upper stopband significantly. The first spurious response appears at 5 GHz, which is 2.58 times away from the central frequency.

an area of 13.1 mm 11.0mm (0.12 g 0.10 g ), which makes a great reduction of 82% in the area. Therefore, our pre-

sented CRLH ring bandpass filter can achieve significant miniaturization and attractive spurious suppression characteristics.

3.2. Dual-Mode Zeroth-Order CRLH Ring Bandpass Filter with Capacitive Stub Loaded

A dual-mode zeroth-order CRLH ring BPF with capacitive stub

and the other dimensions are the same as those in Section 3.1. The measured and simulated S-parameters are plotted Figure 7(b), which are in very good agreement. From the figures, it can be seen that the central frequency is 2.01 GHz with a return loss of 1.6 dB and the 3-dB bandwidth is about 120 MHz. Different against the results shown in Section 3.1, there are two transmission zeros at 1.91 GHz and 2.92 GHz, which can help to improve the performance of both the lower and upper stopbands. Thus, the method to introduce a capacitive stub proposed in [18] is also efficient for our zeroth-order CRLH ring BPF.

4. CONCLUSION

A novel miniaturized dual-mode ring bandpass filter is realized and studied in this article. By using only one resonant-type left-

Figure 6 Dual-mode zeroth-order CRLH ring bandpass filter with only one LH unit: (a) photograph; (b) simulated and measured S-parameters

Figure 7 Dual-mode zeroth-order CRLH ring bandpass filter with capacitive stub loaded: (a) photograph; (b) simulated and measured S- parameters

handed unit with complementary split ring resonator to combine with a microstrip line, the ring configuration can provide two orthogonal zeroth-order resonance modes, and its occupied area can be reduced by about 82% when compared with the conventional one. The filter has the first spurious response 2.58 times away from the central frequency, so its spurious suppression characteristics is also improved. A capacitive stub is introduced to tune the transmission zero frequencies, so its frequency domain characteristics are easier to be controlled. The filter performances are demonstrated by our measured results, with the good agreement obtained when compared with the simulated ones.

REFERENCES

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11.Y.-C. Chiou, J.-T. Kuo, and J.-S. Wu, Miniaturized dual-mode ring resonator bandpass filter with microstrip-to-CPW broadside-cou- pled structure, IEEE Microwave Wireless Compon Lett 18 (2008), 97–99.

12.S. Otto, A. Rennings, C. Caloz, and P. Waldow, Dual mode zeroth order ring resonator with tuning capability and selective mode excitation, Eur Microwave Conf Dig 1 (2005), 4 – 6.

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15.J.D. Baena, J. Bonache, F. Martin, R. Marques, F. Falcone, T. Lopetegi, M.A.G. Laso, J. Garcia, I. Gil, M.F. Portillo, and M. Sorolla, Equivalent-circuit models for split-ring resonators and complementary split-ring resonators coupled to planar transmission lines, IEEE Trans Microwave Theory Tech 53 (2005), 1451–1461.

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©2009 Wiley Periodicals, Inc.

FREQUENCY-INDEPENDENT PERFORMANCE OF ELLIPTIC PROFILE TEM HORNS

Jag Malherbe

Department of Electrical, Electronic and Computer Engineering, University of Pretoria, Pretoria 0002, South Africa; Corresponding author: jagm@up.ac.za

Received 9 July 2008

ABSTRACT: The TEM horn with elliptic E-plane profile has been shown to give extremely wide bandwidth performance as far as VSWR and gain is concerned. In this article, the variation in radiation pattern versus frequency is explored, and it is shown that, dependant on choice, E-plane or H-plane radiation patterns that are virtually independent of frequency can be obtained. © 2009 Wiley Periodicals, Inc. Microwave Opt Technol Lett 51: 607– 612, 2009; Published online in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.24109

Key words: TEM horn; elliptic profile; frequency-independent properties

1. INTRODUCTION

The so-called TEM horn has proven to be extremely popular for the realization of very wide bandwidth performance, where gain and input VSWR are concerned. Various approaches to realizing the desired wide bandwidths are employed, making use variously of impedance and dimensional tapers [1– 6]. In all cases, the emphasis is on obtaining wideband performance, mostly for impedance. The analytical description of the TEM horn is extremely limited, with the exception of [7], where the characteristic impedance of the radial lines that constitute the TEM horns on a point-by-point basis is derived. Thus, to date approaches to design are limited to the realization of the best impedance match, making use of the information of [7], in combination with the various tapers. Little or no effort is made to present information as to the effect of dimensions on parameters such as gain or radiation pattern as a function of frequency.

Recently, the application of an elliptic function to describe the plate separation was introduced [4 – 6], and it was shown that extremely wideband performance could be achieved. However, the radiation pattern properties were not considered per se. It was shown in [6] that very good agreement could be achieved between measured values on a physically constructed horn, and numerical analysis by means of the commercial software FEKO© [8] for a number of variables.

In this article, the validity of the numerical analysis employed in the calculation of horn properties is re-established by extensive comparison of additional measured and calculated

Figure 1 Horn dimensions (mm) [6]. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com]

Figure 2 Unfolded horn plate (mm). [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com]