диафрагмированные волноводные фильтры / ac443111-8240-4f26-ad1f-f50c5d2a9d5f

results indicate that the effect of mutual coupling in E-plane is more severe than in H-plane for the same center-to-center separation of the array elements.

4. CONCLUSIONS

A printed antenna comprising the series-fed dipoles and the CB- CPW-to-CPS balun with broad bandwidth has been investigated experimentally and theoretically. The end-fire radiation patterns with a gain of about 4.5–9.1 dBi have also been measured within the operating frequency range. Mutual couplings of two identical series-fed dipole antennas in face-to-face as well as side-by-side configurations were measured and found to vary significantly with frequency and antenna separation. The signal correlation coefficient obtained form S-parameters of the two-antenna array is lower than 0.04 in the operating band. The presented method can be utilized in array systems for the compensation or exploitation of mutual coupling. The proposed antenna is applicable for the highgain multiantenna system with the advantages of broad bandwidth, low cost, and easy fabrication.

ACKNOWLEDGMENTS

This work was supported in part by the National Science Council of Taiwan, R.O.C., under Grant NSC 95–2221-E-027– 018, Grant 95–2221-E-027– 031, and Grant 95–2752-E-002– 002-PAE.

REFERENCES

1.G. Kumar and K.P. Ray, Broadband microstrip antenns, Artech House, Norwood, MA, 2003.

2.R.S. Elliott, Antenna theory and design, Prentice-Hall, Englewood Cliffs, NJ, 1981.

3.M.K. Ozdemir, H. Arslan, and E. Arvas, Mutual coupling effect in multiantenna wireless communication systems, IEEE GLOBECOM 2 (2003), 829 – 833.

4.S.-G. Mao, C.-T. Hwang, R.-B. Wu, and C.-H. Chen, Analysis of coplanar waveguide-to-coplanar stripline transitions, IEEE Trans Microwave Theory Tech 48 (2000), 23–29.

5.F. Tefiku and C.A. Grimes, Design of broad-band and dual-band antennas comprised of series-fed printed-strip dipole pairs, IEEE Trans Antennas Propag 48 (2000), 895–900.

6.S.-G. Mao and S.-L. Chen, Broadband series-fed printed dipole arrays with conductor-backed coplanar waveguide-to-coplanar stripline transitions, IEEE Antennas Propag Int Symp A 3 (2005), 565–568.

7.S. Blanch, J. Romeu, and I. Corbella, Exact representation of antenna system diversity performance from input parameter description, Electron Lett 39 (2003), 705–707.

© 2007 Wiley Periodicals, Inc.

GAIN ESTIMATION BY CONVERGENCE OF ACTIVE ELEMENT PATTERN FOR E-PLANE NOTCH PHASED ARRAY ANTENNA

Junyeon Kim,1 Joonho So,1 Won Jang,1 and Changyul Cheon2

1 TRC Microwave Division, Agency for Defense Development, Seoul, South Korea

2 Department of Electrical and Computer Engineering, University of Seoul, Seoul, South Korea

Received 26 September 2006

ABSTRACT: The beam steering gain of E-plane notch phased array antenna is estimated using the converged active element pattern (AEP) and the directivity of an isotropic linear array. To get the gain for all

scan angles, the authors first investigated the convergence of an AEP with the number of finite notch array at the center element. And then, the beam steering gain can be calculated from multiplying the converged AEP by the directivity of an isotropic linear array. To show the usability of this proposed method, a 10-element E-plane notch array antenna has been designed and fabricated. The measured AEP of E-plane notch antenna has a good agreement with the converged AEP. © 2007 Wiley Periodicals, Inc. Microwave Opt Technol Lett 49: 1047–1049, 2007; Published online in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop. 22330

Key words: active element pattern; active reflection coefficient; E-plane notch phased array antenna

1. INTRODUCTION

A notch phased array antenna is a good candidate for a wideband phased array antenna and has been used in wide applications [1–3]. In designing the phased array antenna, an accurate prediction of the gain at each beam scan angle is quite important. To predict the gain of the phased array antenna, a method utilizing an active element pattern (AEP) has been developed and used widely [4]. The AEP, which is a performance index for phased array antenna, represents a variation of the gain with scan angle of the array elements which are fully excited to illuminate. One can obtain the AEP by measuring pattern of array antenna or by computing with numerical or theoretical techniques. In designing state, the computation of the AEP using numerical techniques is much attractive because of its cost and flexibility.

In recent years, many researchers have designed the array antenna using numerical computations for a unit cell of the array with periodic boundary conditions by summing the Floquet modes [1, 2], or with the numerical waveguide simulators [3]. In these works, the active reflection coefficient (ARC), or the active impedance, was optimized to obtain the gain of the AEP using simple approximated equation in [4]. The disadvantage of these methods is that they give inaccurate results in spite of a large amount of computational cost because the ARCs at all scan angles should be computed with the assumption that the array is infinite.

In this letter, we proposed the method for estimating the gain of a linear phased array antenna at all scan angles with less time consumption. First, we have investigated the converging characteristics of the AEP as changing the size of finite array by using numerical full wave analysis, and determined the size of finite array which is appropriate to represent the AEP of a large but finite number of array elements. After determining the converged AEP, the gain of a linear phased array antenna with all scan angles is obtained by multiplying the converged AEP by the directivity of an isotropic linear array.

To show the usability of the proposed method, the gains of E-plane notch phased array antenna at each scan angle are calculated in various ways including our method and compared with measured data. And the computing time for estimating the gain is also compared.

2. GAIN ESTIMATION OF LINEAR ARRAY ANTENNA

In classical analysis, the directivity of an ideal linear array at the angle of bore sight, 0, is given by

where N is the number of elements of the linear array, k is the free space propagation constant and d is distance between adjacent

Figure 1 Convergence characteristic of the AEP of E-plane notch phased array antenna. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com]

elements, and 0 is the angle at bore sight [5]. There is a difference between the real gain and the calculated gain by Eq. (1) since mutual couplings between elements which are not counted in Eq.

(1) exist in real array system. And using Eq. (1), we cannot get the gain of all scan angles. So, to consider the radiating mutual couplings in gain, one may use pattern multiplication for gain as

where GPAA denotes the gain of a linear phased array antenna including mutual couplings, GAEP is the gain of the AEP that contains all effects of radiating mutual couplings in the array, is a scan angle, and G 0 is the directivity of an ideal linear array at the angle of bore sight.

To estimate the gain, GPAA with all scan angles, , the accurate AEP has to be obtained first. It is also a difficult task to

solve the whole array to acquire the AEP since it requires huge time and memory for the large number of elements. But if we find the small finite array to represent the AEP of the infinite array, we save a lot of time and memory to solve.

For example, in this article, the AEPs of the E-plane notch array are computed according to the number of elements up to ten. Figure 1 shows the converged characteristic of the AEP for different numbers of elements obtained by using HFSS commercial software. As it is shown in the figure, the gain of the AEP converges as the number of elements increases, especially in the range of scanning angle, that is within the angle of 45°. According to the figure, the gain of the AEP for five-element arrays is enough to approximate that of larger arrays within the scanning volume. So, we used the converged gain of the AEP that were

obtained using numerical technique for five-element E-plane notch

array antenna to calculate the whole gain, GPAA , of Eq. (2) for E-plane notch array antenna. With this approximation, one can

obtain more accurate gain with less time consumption compared with the previously available methods.

3. ESTIMATED GAIN FOR E-PLANE NOTCH PHASED ARRAY ANTENNA

To show the usability of the proposed method, a 10-element E-plane notch phased array antenna is designed [6] and fabricated. Figure 2 shows a 10-element E-plane notch phased array antenna. To evaluate the performance of the fabricated E-plane notch phased array antenna, the measured ARC and the simulated ARC at the center element in E-plane notch phased array antenna are presented in Figure 3. The figure shows that the ARC is low enough in more than one octave frequency range within a scan

Figure 2 E-plane notch phased array antenna. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley. com]

Figure 3 (a) The simulated ARC at center element; (b) The measured ARC at center element. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com]

Figure 4 The gain of the AEP of E-plane notch phased array and the beam steering gain. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com]

volume of 45°. And the measured ARC also shows a good agreement with the simulation result.

Using the proposed method, the gain of E-plane notch phased array antenna is calculated. In Figure 4, the gain of the AEP in proposed method and the measured AEP are shown respectively and also compared with the approximated gain of the AEP using the simulated ARC from unit cell analysis and the results of the measured ARC in [4]. Through the simulation of the fully excited array for the beam steering angle ( 45°, 30°, 15°, and 0°), we can see the gain of the AEP in proposed method has a similar behavior to the beam steering gain.

In the respect of time consuming, computation time for E-plane phased array antenna using proposed method takes 1 hour (h) 31 min (m) 50 seconds (s) using HFSS commercial software, but computation time for unit cell analysis at all scan angles and using approximated equation with the ARCs takes 2 h 22 m 30 s with a computer (Pentium4 1.6 GHz, 1 G RAM). Therefore the proposed method allows a significant reduction (almost 50 m) of computing time with respect to method in [4].

4. CONCLUSION

In this letter, the new method for beam steering gain estimation of linear phased array antenna is proposed. As an example, the E-plane notch phased array antenna is designed and the converged result of the AEP and the minimum number of array having the characteristics similar to the AEP of infinite array are presented. Considering the converged gain of the AEP and the directivity of ideal elements, we can easily estimate the beam steering gain of the finite E-plane notch phased array antenna without calculation or measurement of the ARC.

The proposed method permits the prediction of the gain variation with the beam scan angle of a linear phased array and can be applicable to planar phased array antenna.

REFERENCES

1.J. Shin and D.H. Schaubert, A parameter study of Stripline-Fed Vivaldi notch antenna arrays, IEEE Trans Antennas Propagat 47 (1999).

2.H. Holter, Element for wide-band and very wide-angle phased arrays,

Antenna Propagat Soc, 2001 IEEE Int Symp 2 (2001), 440 – 443.

3. M. Kragalott, W.R. Pickles, and M.S. Kluskens, Design of a 5:1

bandwidth stripline notch array from FDTD analysis, IEEE Trans Antennas Propagat 48 (2000).

4.D.M. Pozar, Active element pattern, IEEE Trans Antennas Propagat 42 (1994).

5.W.L. Stutzman and G.A. Thiele, Antenna theory and design, Wiley, New York, 1981.

6.J.-Y. Kim, J.-H. So, J.-S. Lim, and Chang-Yul Cheon, Design of wide-band/wide-scan E-plane notch phased array considering active element pattern, Antenna Propagat Soc, 2003 IEEE Int Symp 2 (2003), 476 – 479.

© 2007 Wiley Periodicals, Inc.

ULTRA-WIDEBAND BANDPASS FILTER WITH A COMPACT TWO-LAYERED STRUCTURE

Xinan Qu,1 Shun-Shi Zhong,1 and Jie Liu2

1 School of Communication and Information Engineering, Shanghai University, Shanghai 200072, People’s Republic of China

2 Powerwave Technologies Inc., Shanghai 200331, People’s Republic of China

Received 29 September 2006

ABSTRACT: A novel ultra-wideband (UWB) bandpass filter with a compact size is proposed. By using a two-layered substrate configuration and an H-shaped slot on the ground, a vertical-coupled microstrip resonator structure is formed. Three resonant peaks of the proposed structure are extracted around the lower end, the center, and the upper end of the UWB passband. The compact two-layered filter has an overall length of about a half guided wavelength at the center UWB frequency and exhibits promising performance for UWB applications. © 2007 Wiley Periodicals, Inc. Microwave Opt Technol Lett 49: 1049 –1051, 2007; Published online in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.22351

Key words: ultra-wideband (UWB); bandpass filter; two-layered

1. INTRODUCTION

Since the Federal Communications Commission (FCC) released the unlicensed ultra-wideband (UWB) frequencies (3.1–10.6 GHz) for commercial use in February 2002, the passive filters for UWB applications have been receiving great interests [1]. Since many techniques widely used for broadband microwave filter design cannot be directly applied to the filters with a 110% fractional bandwidth, some new structures have been proposed in the last several years. A microstrip ring filter with a passband from 3 to 10 GHz was presented in Ref. 2. The filter suffered unexpected passbands below 3 GHz and near 13 GHz. In Ref. 3, a full band UWB filter was derived from high pass filters but the reported insertion loss around 10 GHz was about 4.7 dB. The microstrip- to-CPW transition with a frequency-dependent characteristic was analyzed in Ref. 4 and the UWB bandpass filters using this configuration showed good in-band performances [5–7].

In this letter, a two-layered structure is selected to form a novel compact UWB bandpass filter. By using a two-layered structure, the overall length of the filter has been decreased to about a half guided wavelength at the center frequency of the operating band. With an H-shaped slot on the ground, the filter achieves tightened vertical coupling and exhibits a UWB bandpass characteristic. The simulation of the proposed filter is carried out on HFSS, which is based on the finite element method (FEM). The design of the UWB filter is presented and the results of test filter are discussed.