диафрагмированные волноводные фильтры / b64e3d4d-69b5-4628-967e-ede99b187db7

Abstract—Two compact planar magic-T structures implemented on the double-sided parallel-strip line (DSPSL) and the slotline coupling are proposed and demonstrated. Due to a metallic ground inserted into the middle of the substrate for the double-sided par- allel-strip line, the E-plane DSPSL power divider thus shows an intrinsic 180 out-of-phase characteristic between the two input/ output ports. A transition between slotline and DSPSL is utilized to realize the H-plane input/output port of the magic-T. The equivalent circuits are derived for the design of the proposed planar multilayer magic-T. Good agreement between the measured and simulated results indicates that the fractional bandwidths for the two planar magic-T structures are 42.8% and 43.8%, respectively, while the amplitude and phase imbalance are less than 0.22 dB and 1.5.

Index Terms—Double-sided parallel-strip line, equivalent circuit, fractional bandwidth (FBW), planar magic-T, slotline.

I. INTRODUCTION

MAGIC-T is a four-port microwave junction, actually Aa 180hybrid in the ideal case, in which there are a sum (H) port and a difference (E) port that allow in-phase or out-of-phase incident signals from two input/output ports, respectively, where the H-plane and E-plane junctions are combined together to realize equal-power transmission between the two input/output ports. Due to these excellent properties, the magic-T structures have been widely used as important elements in correlation frequency discriminators, balanced mixers, receivers, and four-port circulators [1] in radar and communication systems. However, the 3-D structure prevents their wide applications in planar integrated microwave circuits. In recent years, many efforts have been focused on the implementation of planar magic-Ts.

Several broadband planar magic-Ts have been proposed [2]–[12], including those implemented with slotline and coplanar waveguide (CPW), those combining microstrip and substrate integrated waveguide (SIW), and several others. The

Manuscript received March 10, 2010; revised June 16, 2010; accepted July 19, 2010. Date of publication October 07, 2010; date of current version November 12, 2010. This work was supported in part by the Shenzhen Science and Technology Planning Project for the Establishment of Key Laboratory in 2009 through Project CXB200903090021A and by the Science and Technology Development Fund of Macao SAR under Grant 020/2009/A1.

W. J. Feng and Q. Xue are with the State Key Laboratory of Millimeter Waves (Hong Kong), City University of Hong Kong, Hong Kong, China (e-mail: eeqxue@cityu.edu.hk; fwj1985159@yahoo.com.cn).

W. J. Feng and W. Q. Che are with the Department of Communication Engineering, Nanjing University of Science and Technology, 210094 Nanjing, China (e-mail: yeeren_che@yahoo.com.cn).

Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TMTT.2010.2078312

symmetric magic-Ts using the slotline structure have a broadband power division response; however, the magic-T using the CPW–slotline transition requires air bridges to prevent the undesired modes, causing fabrication complexity and higher cost [2], [3]. Moreover, to realize impedance matching at four ports of magic-Ts with the microstrip–slotline transition, one inclining structure is required, which increases the complexity of circuit design. In addition, the transmission line between the transition and T-junction are a quarter/half wavelength long, which occupies relatively a large area of the magic-T [4]–[9].

With the advent of advanced technologies, the multilayered microwave integrated circuits have become popular for microwave components [13]–[16] using conventional fabrication technology. The double-sided parallel-strip line (DSPSL), as one kind of balanced transmission lines, is quite useful and convenient for the multilayered microwave integrated circuits designs. Compared with other balanced transmission lines, such as the coplanar stripline, DSPSL has important advantages of easy realization of low and high characteristic impedance, simple circuit structures of wideband transitions, and good performance of balanced microwave components [17]–[19]. In addition, it can be analyzed easily using image approach, as found in the literatures [20], [21]. The transitions between different transmission lines and DSPSL have been illustrated in [17]–[19], [22], [23]. However, fewer papers have been published about the applications of the transition between slotline and DSPSL.

To achieve simpler design and further size reduction of the conventional slotline magic-T structures, this paper presents two compact magic-Ts consisting of a transition between slotline and DSPSL and an E-plane DSPSL power divider [18], [23]. Due to the three-layer coupling structures, fewer quarter-/half- wavelength transmission lines for impedance matching at each port are required. In addition, two simplified equivalent circuits are developed to provide a conceptual interpretation of the structure operation. All the structures are simulated with Ansoft High Frequency Structure Simulator (HFSS) v.10.0 and constructed into the dielectric substrate with and mm.

II.DESIGN OF THE TWO PLANAR MAGIC-TS

A. Proposed Two Planar Double-Sided Parallel-Strip Line Magic-Ts

Figs. 1 and 2 illustrate the 3-D and top views of the two proposed planar DSPSL magic-Ts with a slotline etched on the inserted metallic ground of the middle substrate. Both structures consist of three layers, and the shadow stands for the metallization parts on different layers. Ports 1 and 4 are the difference

(E) and sum (H) ports of the magic-T, respectively, and two

Fig. 1. Proposed planar DSPSL magic-T with end coupling. (a) 3-D view. (b) Top view.

Fig. 2. Proposed planar DSPSL magic-T with center coupling. (a) 3-D view.

(b) Top view.

power-division ports (port 2, 3) are located symmetrically on the top and bottom of the structure. The difference of the proposed two magic-T structures lies in the coupling transition between the slotline and DSPSL. End coupling of two steppedimpedance open stubs are utilized to realize the signal coupling to arm H (port 4) in the first magic-T. Meanwhile, due to the intrinsic 180 phase difference of the E-plane DSPSL power divider, an improved coupling transition between slotline and coupling transition without stepped-impedance open stubs, for center coupling, are employed for energy coupling to arm H (port 4) in the second magic-T.

Fig. 3(a) and (b) describes the electric fields at the A–A plane and B–B plane, respectively. Due to the metallic ground in the middle of the substrate, 180 phase difference is achieved between ports 2 and 3 when the signal is injected from port 1, an equal out-of-phase power division can thus be realized between ports 2 and 3. In addition, due to the slotline etched on the middle metallic ground, the electric field is coupled symmetrically from the middle-layer slotline to the microstrip lines on the top and bottom layers. In this way, an equal in-phase power division is realized when the signal is transmitted from port 4 to ports 2 and 3. Due to the 180 phase difference for ports 2 and 3, when the signal is transmitted from port 1 to ports 2 and

Fig. 3. (a) Electric field distribution at the A–A plane. (b) Electric field distribution at the B–B plane.

3, the symmetrical C–C plane becomes a virtual electric wall, and port 4 is then isolated. Similarly, a virtual magnetic wall is induced at the symmetrical C–C plane when the signal is transmitted from port 4 to ports 2 and 3, and port 1 is thus isolated because of the in-phase characteristic for ports 2 and 3. Moreover, the size of the magic-T is further reduced due to the three-layer structure.

B. Circuits for End-Coupling Magic-T and Center-Coupling Magic-T

Some simplified equivalent circuits for microstrip-slotline magic-Ts have been illustrated in [5], [8], [9], and [24]. Here,

Fig. 4. Complete circuit models of the two planar magic-Ts at center operating frequency. (a) End coupling. (b) Center coupling.

an equivalent circuit similar to that in [8] is used to provide a conceptual understanding of the two planar DSPSL magic-Ts. Fig. 4 illustrates the circuits for the two planar magic-Ts, approximately representing the frequency responses of the magic-Ts around the center frequency .

In the circuit models, an ideal 1:1 transformer is introduced to model the 180 phase difference that is characteristic of the equal power division transmitting from port 1 to ports 2 and 3. Meanwhile, for the microstrip–slotline transition models from port 4 to ports 2 and 3, a conventional simplified circuit [5] is introduced, where the microstrip–slotline impedance-matching performance depends on the transformer ratio of the ideal transformers, and is quite related to the substrate thickness, the transmission-line characteristic impedance, as well as the microstrip–slotline physical alignment [5]. The definition of is

(1)

The required ratio can be achieved by adjusting , , and , the inclination angle between the microstrip line and slotline. It can be conveniently calculated by the method in [25]–[27] as

(2)

In order to acquire a broadband mode conversion for the microstrip–slotline transition, a microstrip stepped-impedance open stub using two transmission lines and with electrical lengths and , and a slotline radial stub with electrical length , are used to realize the coupling between

Fig. 5. Simulated magnitude response of the end-coupling magic-T.

(a) Out-of-phase. (b) In-phase.

the microstrip and slotline [24]. The characteristic impedances of the slotline can be calculated analytically [5]. Considering the effect of the radiation loss of the slotline and the limitation of fabrication precision, where we choose the substrate with dielectric constant 2.65 and thickness of 0.5 mm, the slotline

Fig. 6. Simulated phase response of the magic-T for the end-coupling magic-T.

(a) Out-of-phase. (b) In-phase.

Fig. 7. Simulated magnitude response of the magic-T for the center-coupling magic-T. (a) Out-of-phase. (b) In-phase.

width is then 0.2 mm, corresponding to 150 . Because the impedance of the microstrip is 50 , then 0.58. The calculated parameters for the two equivalent circuits are listed in Table I, along with the corresponding optimization dimensions for the two magic-Ts in Figs. 1 and 2.

C. Simulated Results for Two DSPSL Magic-T Structures

The simulated magnitude and phase responses of circuits (simulated with Ansoft Designer v3.0) and structures (simulated with Ansoft HFSS v.10.0) for the two DSPSL magic-Ts are illustrated in Figs. 5–8. The whole length of the open stub connected to ports 2 and 3 in Fig. 1 is . Due to the influence of a microstrip stepped-impedance open

feature of the stepped-impedance resonator (SIR) [27]. In addition, the phase distortions at and can be observed in the simulated phase in Fig. 6(a), due to the existence of the transmission zeros of the open-circuited lines.

Comparing the simulated magnitude in Fig. 5(a) for the magic-T with end coupling with the center-coupling magic-T of Fig. 7(a), we may see that, due to a lack of the open-stub effect for the power transmission from port 1 to ports 2 and 3 of the center-coupling magic-T, a wider band for the equal

power divider can be realized from the DSPSL to the microstrip line in the magic-T with center coupling. In addition, due to effects of the radiation loss of the slotline and the impedance match between the slotline and microstrip line in the practical magic-T structures in Figs. 1 and 2, narrower equal in-phase power division bands are found for both DSPSL magic-Ts, as observed from the simulated results of the slotline–microstrip transition for end-coupling magic-T in Fig. 5(b) and for the center-coupling magic-T in Fig. 7(b).

In Fig. 5, the insertion loss for out-of-phase and in-phase for the end-coupling magic-T is less than 3.65 and 3.7 dB (5.6–8.6 GHz). For the center-coupling magic-T, the insertion loss for out-of-phase is less than 3.6 dB from 2 to 10 GHz and that for in-phase is less than 3.68 dB from 5.1 to 8 GHz, as shown in Fig. 7.

III. MEASURED RESULTS AND DISCUSSIONS

Two prototypes of the proposed planar DSPSL magic-Ts with a size of 36 mm 28 mm (the end-coupling magic-T) and 30 mm 36 mm (the center-coupling magic-T) are fabricated and measured. Fig. 9 illustrates the photographs of the two planar DSPSL magic-Ts. We may note that, in the measurement, the central conductor of the SMA connector is connected with the top strip of DSPSL while its outer conductor is connected with the bottom strip of DSPSL, meaning that the two strips function as the grounding mutually.

Fig. 8. Simulated phase response of the magic-T for the center-coupling magic-T. (a) Out-of-phase. (b) In-phase.

Fig. 9. Photographs of the two proposed planar magic-Ts. (a) End-coupling magic-T. (b) Center-coupling magic-T.

However, it must be mentioned that, in this case, a single-ended coaxial line represented by SMA connector is directly connected to the differential line represented by

DSPSL, causing an unbalanced structure. Because of this, some electromagnetic field will occur between the outer conductor of the coaxial line connected to the SMA connector on port 1 and a global ground connected to the outer conductors of SMA

Fig. 11. Measured and simulated results of amplitude and phase imbalances of the end-coupling magic T. (a) Amplitude. (b) Phase.

connectors of the other three ports and thus result in some sensitivity of properties of the designed magic-T to the presence of objects in its vicinity and may cause slight performance deterioration. However, for further practical applications, a transition to microstrip can be introduced to achieve a global ground for the four ports of this magic-T and thus avoid this problem. In addition, a better solution to this problem is currently under study.

Measured results are shown in Figs. 10–13, as well as the simulation results; good agreement is obviously observed. For the magic-T with end coupling, within 5.5–8.5 GHz, the insertion loss for out-of-phase and in-phase is less than 3.75 and 3.71 dB, while the return loss for ports 1 and 4 is greater than 11.2 and 16 dB, respectively. Meanwhile, and are greater than 20 and 30 dB, respectively. The amplitude and phase imbalances for out-of-phase and in-phase are less than 0.2 dB and 1.4. For the center-coupling magic-T, within 5–7.8 GHz, the insertion loss for out-of-phase and in-phase is less than 3.63 dB (2–9.5 GHz) and 3.73 dB, the return loss for ports 1 and 4 are greater than 11.5 dB and 15 dB, while 23 and 41 are greater than 18.5 dB and 32 dB, respectively. The amplitude and phase imbalances for out-of-phase and inphase are less than 0.22 dB and 1.5.

Fig. 12. Measured and simulated results of the center-coupling magic-T.

(a) Out-of-phase. (b) In-phase (c) and . (d) Isolation.

The slight frequency discrepancy and slightly larger insertion loss for these two DSPSL magic-Ts are probably caused by

Fig. 13. Measured and simulated results of amplitude and phase imbalances of the center-coupling magic-T. (a) Amplitude. (b) Phase.

the fabrication inaccuracy and the imprecise alignment of the two substrate boards. In order to realize accurate orientation for the top and bottom layers of the practical magic-Ts, some positioning holes are fabricated to achieve good alignment of these

two substrate layers. However, if lamination process skill can be used in the fabrication, the performances of these two magic-Ts should be better.

In addition, comparisons of measured results for several different magic-T structures are illustrated in Table II. The proposed two DSPSL magic-T structures almost maintain a broadband transmission characteristic similar to the conventional slotline magic-T structures; however, due to the three-layer structure of the two DSPSL magic-Ts, some redundant quarter-/half-wavelength transmission line for impedance matching at each port are not required, and a simple design and further size reduction can thus be realized. Actually, DSPSL is a wideband transmission line; however, the transition between slotline and microstrip affects the bandwidth significantly. In this way, the possible technique to enhance the bandwidth is to design a wideband transition. In addition, narrower slotline with higher impedance can be used to realize better coupling between the slotline and the microstrip and then achieve wider bandwidth, which demands more precise fabrication.

IV. CONCLUSION

In this paper, two compact planar DSPSL magic-Ts implemented by an E-plane DSPSL power divider connected with a simple transition between slotline and DSPSL are designed and analyzed. Due to the metallic ground located in the middle of the structure, the compact E-plane DSPSL power has an intrinsic 180 reverse phase characteristic. The transition between slotline and DSPSL is now introduced to the E-plane DSPSL power divider to realize the wider in-phase transmission band for the two planar DSPSL magic-Ts. Two simplified equivalent circuits are proposed to better understand and design the two planar magic-Ts. Good agreement can be found between the simulated and measured results over the -band. With a simple design and compact structure, this kind of planar magic-T is expected to find wide application in highly integrated microwave circuits of communication and radar systems.

ACKNOWLEDGMENT

The authors would like to thank Dr. X. Y. Zhang, School of Electronic and Information Engineering, South China University of Technology, Guangzhou, China, for his valuable discussions and experimental help during this work. In addition, the authors would like to thank the editors and reviewers of this paper for their valuable comments and suggestions, which have greatly improved the quality of this paper.

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Wen Jie Feng (M’10) was born in Shangqiu, Henan Province, China, in 1985. He received the B.Sc. degree from the First Aeronautic College of the Airforce, Xinyang, Henan Province, China, in 2008, and the M.Sc. degree from the Nanjing University of Science and Technology (NUST), Nanjing, China, in 2010, where he is currently working toward the Ph.D. degree.

From November 2009 to February 2010, he was a Research Assistant with the City University of Hong Kong, Hong Kong. His research interests include mi-

crostrip circuits and microwave passive substrate integrated components and systems.

Mr. Feng is a reviewer for the IEEE MICROWAVE AND

COMPONENTS LETTERS.

Quan Xue (M’02–SM’04) received the B.S., M.S., and Ph.D. degrees in electronic engineering from the University of Electronic Science and Technology of China, Chengdu, China, in 1988, 1990, and 1993, respectively.

In 1993, he joined the University of Electronic Science and Technology of China (UESTC), Chengdu, China, as a Lecturer. He became an Associate Professor in 1995 and a Professor in 1997. From October 1997 to October 1998, he was a Research Associate and then a Research Fellow with the Chinese Univer-

sity of Hong Kong. In 1999, he joined the City University of Hong Kong, Hong Kong, where he is currently a Professor and serves as the Deputy Director of State Key Laboratory (Hong Kong) of Millimeter-waves of China. Since May 2004, he has been the Principal Technological Specialist of the State Integrated Circuit (IC) Design Base, Chengdu, China. He has authored or coauthored over 170 internationally referred papers and over 60 international conference papers. He is an editor of the International Journal of Antennas and Propagation. His current research interests include microwave/millimeter-wave components and subsystems, antenna, microwave monolithic integrated circuits, RF identification, and RF integrated circuits.

Dr. Xue is an Associate Editor of the IEEE TRANSACTIONS ON MICROWAVE

THEORY AND TECHNIQUES and the IEEE TRANSACTIONS ON INDUSTRIAL

ELECTRONICS. He is the Coordinator of Region 10 of the IEEE Microwave Theory and Techniques Society (MTT-S) AdCom.