диафрагмированные волноводные фильтры / d0d6be56-2b7b-4b34-829a-71672f88b751

Abstract—This paper proposes a compact reconfigurable (bandstop/bandpass) and frequency-tunable structure based on S-shaped split ring resonators (S-SRRs). It is known that an S-SRR coupled to a coplanar waveguide (CPW) provides a stopband in the transmission characteristic of the line. It is shown here that this behaviour of the S-SRR can be switched between fundamental resonance and second harmonic response by introduction of a PIN diode in the center segment of the S-SRR. Alternatively, if the S-SRR is loaded with a varactor diode instead of a switch, the frequency of the stopband can be continuously tuned from the S-SRRs fundamental resonance frequency to its second harmonic. Furthermore, it is shown that if a pair of shunt PIN diodes are introduced across the slots of the host CPW, the structure can be reconfigured from a bandstop to a bandpass structure. Thus, the proposed resonator structure can be used as the building block of reconfigurable (bandstop/bandpass) filters with tunable operating frequency. Finally, in order to demonstrate a practical application of the proposed structure, an ultrawideband antenna with a tunable band-notch is designed and experimentally validated.

Index Terms—Reconfigurable antennas, S-shaped split ring resonator (S-SRR), ultrawideband antenna.

I. INTRODUCTION

The rapid proliferation of wireless communication systems has created a need for advanced microwave devices such as reconfigurable or tunable filters and antennas. These types of devices minimize the need to undergo expensive costs associated with the re-installation of wireless infrastructures, since a change in the frequency, bandwidth, or other characteristics of the hardware can be achieved through electronic/mechanical reconfiguration or tuning [1]. Reconfigurable and tunable devices also provide the required hardware for a more efficient management and use of the spectrum through the concepts

This work has been supported by the Australian Research Council under project DP120100661, by MINECO (Spain) under project TEC2013-40600- R, by Generalitat de Catalunya under project 2014SGR-157, and by FEDER Funds. Ferran Martn acknowledges also the suport of ICREA.

Ali K. Horestani is with the School of Electrical & Electronic Engineering, The University of Adelaide, Adelaide, SA 5005, Australia and also with the Ministry of Science, Research and Technology, Tehran, Iran. (e-mail: Ali.K.Horestani@ieee.org ).

Z. Shaterian and C. Fumeaux are with the School of Electrical & Electronic Engineering, The University of Adelaide, Adelaide, SA 5005, Australia.

Jordi Naqui, and Ferran Mart´ın are with GEMMA/CIMITEC, Departament d’Enginyeria Electronica,´ Universitat Autonoma´ de Barcelona, 08193 Bellaterra, Spain (e-mail: Ferran.Martin@uab.es).

Copyright (c) 2012 IEEE. Personal use of this material is permitted. However, permission to use this material for any other purposes must be obtained from the IEEE by sending a request to pubs-permissions@ieee.org.

of dynamic spectrum access and cognitive radio [2], [3]. Application of reconfigurable and tunable devices also permits the time sharing of hardware, which in turn leads to mass and size reduction of the communication systems. This is an important aspect in portable devices and of significant importance in satellite communication systems [4]. To address this demand different reconfigurable and/or tunable antennas based on various techniques have been proposed. For instance, slot resonators and open-loop resonators loaded with PIN or varactor diodes have been used to achieve UWB antennas with a tunable notch band [5]–[10]. Alternatively, to achieve more compact reconfigurable structures with simpler biasing circuits RF-MEMS devices were integrated with different antenna structures to achieve antennas with switchable operating frequency or UWB antennas with on-demand WLAN rejection [11], [12].

Another important aspect towards reducing the overall cost, mass, and size of communication systems is the miniaturization of the microwave components including microwave resonators. In particular, a special attention has been devoted to the miniaturization of metamaterial resonators, which is the key in their application for the implementation of compact planar microwave components such as filters [13], [14], sensors [15], [16], and antennas [17]–[20]. To further reduce the electrical size of metamaterial-inspired resonators, several strategies have been reported [21]–[23]. Among them, the transmission characteristics of S-shaped SRRs (S-SRRs) [24]– [27] has been studied by some of the authors, and it was shown that a high level of miniaturization can be achieved when an S- SRR is excited by the contra-directional magnetic fluxes from the slots of a coplanar waveguide (CPW) [25], [27].

This paper tackles both above mentioned challenges, i.e. reconfigurability of the antenna and miniaturization of the tunable part, by proposing a reconfigurable and tunable S- shaped resonator. It is shown that by introducing PIN/varactor diodes in the design of the S-SRR a compact structure with switchable/tunable operating frequency can be achieved. Further, by adding PIN diodes strategically placed on the CPW loaded with S-SRRs, the bandstop behavior of the overall structure can be reconfigured to a bandpass behavior. The concept proposed in this study leads to a compact reconfigurable and tunable structure that could be applied to a wide range of microwave components such as bandstop and bandpass filters, antennas, and frequency selective surfaces. The novelty of the structure lies in the demonstration of a tunable structure

combining the following features:

–Single-layer implementation (on substrate backside) uniquely adapted to CPW technology.

–Potential to achieve reconfigurability between bandpass and bandstop operation.

–Wide frequency tunability of nearly one octave arising from a change in topology of the resonator, from a pair of separate SRRs to a single S-SRR.

–Relative simplicity of the biasing structure allowing in principle both reconfigurability and frequency tunability.

–Compactness resulting in pronounced effect on resonance and minimally invasive operation outside resonance.

As illustration, in order to demonstrate application of the proposed structure, a design of ultra-wideband (UWB) antenna with tunable notch bands to minimize the interference with WiMax or WLAN systems is presented here.

This paper is organized as follows. Section II starts with a short review on the principle of excitation of the S-SRRs by the fundamental mode (even mode) of a CPW. It then demonstrates how to create bandstop/bandpass reconfigurable structures with switchable and tunable operating frequency. In order to demonstrate a potential application of the proposed structure, a design of a UWB antenna with a tunable notch band is proposed in Section III, with successful experimental validation. Finally, the main conclusions of the study are highlighted in Section IV.

II. RECONFIGURABLE AND TUNABLE S-SHAPED SRRS IN

CPW TECHNOLOGY

For the readers convenience, the first part of this section shortly reviews the working principle and excitation method of the fundamental resonance of the S-SRRs in CPW technology [25]. The second subsection demonstrates that a stopband with switchable/tunable frequency can be achieved in the transmission characteristic of a CPW if it is coupled to an S-SRR that is loaded with a PIN/varactor diode. The third part of this section is further developing the concept by proposing a reconfigurable (in terms of bandstop/bandpass behavior) and frequency switchable/tunable structure, which is experimentally demonstrated in the final part of this section.

A. S-shaped SRRs: Working Principle and Excitation of the Fundamental Resonance in CPW Technology

The typical configuration of an SRR-loaded CPW for filter applications is illustrated in Fig. 1 (a). The structure consists of a pair of rectangular single-ring SRRs etched on the back side of the substrate and centered underneath the slots of the CPW. Note that, as shown in the figure, the magnetic fields of the CPW’s slots are contra-directional. However, since each slot is loaded with a separate SRR, a non-zero net magnetic flux passes through the surface of each SRR. The magnetic coupling between the CPW and the pair of SRRs excites the resonators, resulting in a stopband in the transmission characteristic of the CPW [28], [29]. In contrast, as demonstrated in [16], [30], [31], if the CPW is loaded with only one SRR that is centered and aligned with the symmetry plane of the CPW (not shown in the figure), the net magnetic

Fig. 1. Bottom view of coplanar waveguides (shown in orange color) loaded with (a) a pair of single-ring SRRs (shown in yellow) etched on the backside of the substrate under the CPW slots, (b) an S-SRR (shown in yellow). Note that, the induced currents by the contra-directional magnetic fields of the CPW’s slots in both rings of the S-SRR are in same direction, resulting in the excitation of the fundamental resonance of the S-SRR.

flux passing through the SRR surface is zero. Thus, in such configuration, the SRR is not excited by the CPW, and does not create a spectral notch in the transmission characteristics of the CPW.

The application of S-SRRs coupled to transmission lines was recently studied. In particular, it was demonstrated in [25] that because of its twisted layout, an S-SRR can take advantage of the contra-directional magnetic fields of the two slots of the CPW. This is because, as shown in Fig. 1(b), the currents induced on both rings of the S-SRR in this configuration are in the same direction. Thus, the S-structure can be excited by the CPW fields at its fundamental resonance frequency. Figure 2 compares the simulated current distribution of the structures shown in Fig. 1(a) and (b) at their fundamental resonances. Simulations are carried out using the 3D electromagnetic solver Ansys HFSS. In both simulations, a 0.257 mm thick Rogers Duroid 5880 material with relative permittivity of 2.2 and 35 m thick copper metalization is used as the substrate. The width w of the CPW’s central strip is 3 mm and the slots’ width is s = 0:1 mm, which correspond to a 50 characteristic impedance. The dimensions (defined in Fig. 2) of the pair of single-ring SRRs are a = 7:5 mm, b = 2 mm, c = 0:5 mm, g = 0:5 mm, and the lateral space between the two loops is d = 1 mm. The dimensions of the S-SRR are exactly the same. It is important to note that the equivalent inductance of the S-SRR is much higher than that of the pair of the conventional SRRs with the same total size. Thus, the resonance frequency of the S-SRR is almost half of the resonance frequency of the pair of SRRs. In other words, the S-SRR offers a high level of miniaturization compared to the pair of conventional SRRs [25].

B. S-SRR with Switchable or Tunable Resonance Frequency in CPW Technology

Let us now focus on developing resonators with switchable or tunable resonance frequency based on S-SRRs in CPW technology. As mentioned in the previous subsection, due to the much larger equivalent inductance of an S-SRR compared to that of a pair of conventional SRRs, the fundamental resonance frequency of the S-SRR is roughly half of the

Fig. 2. Simulated current distribution of (a) a pair of single-ring SRRs, and (b) an S-SRR excited at their fundamental resonance frequencies by the fundamental mode (even mode) of a CPW.

Vbias

D1

Fig. 3. A CPW loaded with a switchable S-SRR. The switchable S-SRR is constructed by diagonally connecting two single-ring SRRs through the PIN diode D1. The structure can act as a tunable resonator if the two single-ring SRRs are connected through a varactor diode instead of the PIN diode.The voltage to switch the PIN diodes is applied through a pair of bias tees.

fundamental resonance frequency of a pair of conventional SRRs with the same size. We also note that, as demonstrated in Fig. 2, an S-SRR can be constructed by diagonally connecting the two single-ring SRRs. This suggests that a pair of singlering resonators can be readily reconfigured to form an S-SRR by diagonally connecting the two single-ring SRRs through an RF switch, such as a PIN diode, as shown in Fig. 3. The PIN diode can be switched ON or OFF by applying an appropriate DC bias voltage through a pair of RF-isolating inductors connected to both sides of the S-SRR. To validate the concept, a CPW loaded with such a frequency-switchable resonator is simulated using HFSS. Lumped-element boundary conditions with nominal series resistance (R = 1 ), and total capacitance (C = 0:14 pF) of an available PIN diode are used for the simulation of the switch in its ON and OFF states, respectively. The substrate material and dimensions, as well as the dimensions of the CPW and the resonators correspond to the simulated structures of the previous subsection. Figure 4 shows the simulated transmission coefficients of the structure for ON and OFF states of the PIN diode. The simulated results show that, when the PIN diode is in OFF state, the two singlering SRRs are disconnected and the structure provides a notch band at f = 5:5 GHz (red dashed line). In the PIN diode’s ON state, however, the two single-rings are diagonally connected, forming an S-SRR. Thus, the notch band is switched to a lower frequency at f = 3:25 GHz (blue solid line). Note that, the ratio of the two frequencies is less than two. This is because in its OFF state the PIN diode is not an ideal open circuit – it shows an OFF-state capacitance C = 0:14 pF.

While the frequency switching capability of the proposed

Fig. 4. Simulated transmission coefficients of the structure of Fig. 3 for both ON (blue solid line) and OFF (red dashed line) states of the PIN diode (D1).

Frequency (GHz)

Fig. 5. Simulated transmission coefficients of the tunable version of the structure of Fig. 3 for different capacitances of the varactor diode D1.

S-SRR structure can be used in various types of devices such as switchable filters, the structure can be further improved to an S-SRR with tunable resonance frequency. To this end, the RF switch (PIN diode) on the S-SRR needs to be replaced with a varactor diode. The layout of the proposed structure is identical to the switchable structure of Fig. 3, except that the PIN diode D1 is now replaced with a varactor diode. In order to change the capacitance of the varactor, the resonator is reverse biased through RF blocking chokes or large resistors.

To validate the principle, the proposed tunable structure is simulated with HFSS. The dimensions of the simulated structure correspond to those of the switchable resonator. For simulation purposes, the nominal resistance and capacitance of the varactor diode are modeled with lumped-element RLC boundary conditions [32]. The simulated transmission coefficients of the structure for various values of the capacitance of the varactor diode are shown in Fig. 5. When the varactor is operated at a high reverse bias voltage, it exhibits its smallest capacitance C = 0:05 pF, corresponding to a relatively high series impedance. In this case, the resonator acts as two separate single-ring SRRs, creating a stopband at about f = 5:7 GHz. In contrast, at zero bias voltage, where the varactor has its highest capacitance C = 2:7 pF, the relatively small diode’s impedance short circuits the two single-ring resonators to form the S-SRR. Thus, the stopband

states between the mentioned extremes, the frequency of the stopband can be tuned about one octave, roughly between the resonance frequency of an S-SRR and that of a pair of SRRs.

C. Reconfigurable Bandstop/Bandpass Structures with Switchable/Tunable S-SRRs

The previous paragraphs have considered a bandstop configuration where an S-SRR at resonance coupled to a CPW can inhibit the propagation of electromagnetic waves along the line. This stop band can be interpreted as the frequency band where the SRR-loaded CPW act as a one-dimensional medium with negative effective permeability. Thus, the bandstop behavior of an SRR-loaded microstrip or CPW line can be switched to a bandpass behavior if the structure is modified to also provide a negative effective permittivity, resulting in a one-dimensional left-handed medium. This can be achieved by introducing a shunt inductive strip between the signal strip and ground plane of the transmission line [25], [26], [29], [33]. Thus, to achieve a reconfigurable device with capability of switching between bandstop and bandpass behavior, as shown in Fig. 6, a pair of shuntly connected PIN diodes D2 and D3 are introduced between the signal strip and the ground planes of the S-SRR-loaded CPW. The PIN diodes can be switched ON or OFF by applying an appropriate DC voltage between the signal strip and the ground planes of the CPW through a bias tee. Thus, when the PIN diodes D2 and D3 are in OFF state the loading resonator inhibits the propagation of the electromagnetic waves along the line, providing a stopband. In contrast, when these PIN diodes are in their ON state, the resonator-loaded CPW shows a bandpass behavior. This feature can be used, in conjunction with the switchablity of the resonance frequency of the S-SRR, to achieve a structure with reconfigurable bandpass/bandstop behavior and switchable frequency. Figure 7 shows the simulated transmission coefficients of the proposed structure for both ON and OFF states of the PIN diode D1 on the S-SRR, while the shuntly connected PIN diodes (D2 and D3) are in ON state. The substrate material and physical dimensions of the simulated structure correspond to those of the simulations in the previous subsection. The simulation results clearly show that when diodes D2 and D3 are ON, the structure exhibits a bandpass behavior with capability of switching between the two resonance frequencies. Note that, in order to achieve optimum insertion loss in the passband, the PIN diodes are aligned with the center point of the S-SRR [34].

A summary of the operating frequencies and bandstop/bandpass behaviors of the proposed reconfigurable and switchable structure for different states of the PIN diodes is given in Table. I.

From what has been so far proposed in the current and previous subsections, it is easily concluded that a reconfigurable (bandstop/bandpass) and frequency tunable structure can be achieved if the PIN diode D1 in the structure of Fig. 6 is replaced with a varactor diode. The simulated transmission coefficients of such reconfigurable and frequency tunable structure for various values of the varactor diode D1 are shown in Fig. 8, for the case where the shuntly connected PIN diodes (D2 and D3) are in ON state.

Fig. 6. A CPW loaded with a switchable/tunable S-SRR as well as a pair of shuntly connected PIN diodes D2 and D3. In their ON state, the pair of PIN diodes between the signal strip and ground planes of the CPW act as a pair shunt inductors, switching the behavior of the structure from bandstop to a bandpass characteristics.

Frequency (GHz)

Fig. 7. Simulated transmission coefficients of the structure of Fig. 6 for both ON and OFF states of the PIN diode D1, while the shuntly connected PIN diodes (D2 and D3) are in ON state.

TABLE I

A SUMMARY OF THE OPERATING FREQUENCIES AND BANDSTOP/BANDPASS BEHAVIORS OF THE PROPOSED STRUCTURE OF

FIG. 3 FOR DIFFERENT STATES OF THE PIN DIODES D1, D2, AND D3.

Frequency (GHz)

Fig. 8. Simulated transmission coefficients of the tunable version (i.e., once the diode D1 is replaced with a varactor diode) of the structure of Fig. 6 for different capacitances of the varactor diode while the shuntly connected PIN diodes (D2 and D3) are in ON state.

Fig. 9. Photographs of the top and bottom views of the fabricated prototype of the proposed reconfigurable and tunable structure. The figure also shows the zoomed pictures of the varactor-loaded resonator, its biasing network, and PIN switches on the CPW side.

D. Experimental validation of reconfigurability and frequency tunability

In order to experimentally validate the concept and simulation results of the proposed reconfigurable and tunable structure, a prototype has been fabricated and measured. Fig. 9 shows photographs of the top and bottom views of the fabricated prototype, which is realized on a 0.257 mm thick Rogers Duroid 5880 material with relative permittivity of 2.2, a loss tangent tan = 0:0009 and 35 m thick copper metalization. The dimensions of the CPW and the loading S-SRR correspond to the simulated structures of the previous subsection. The S-SRR is loaded with a Skyworks [35] SMV1405 varactor, that offers a variable capacitance range extending from 0.63 pF to 2.67 pF (i.e. not covering the whole range of capacitance which is used in the simulation). Also, the CPW line is loaded with a pair of MACOM’s [36] MADP- 042305-130600 PIN diodes. Since the varactor’s bias current amounts to only a few nano Ampere, a pair of 1 M resistors is used to block RF signals while providing an adjustable reverse bias to the varactor diode [32]. Note that, the pair of PIN diodes is placed between the signal strip and the ground planes of the CPW. Thus, they are simply switched ON or OFF by applying an appropriate DC voltage to the SMA connectors through a pair of bias-tees.

The measured transmission coefficients of the prototype for different states of the PIN diodes and various reverse-bias voltages of the varactor diode are depicted in Figs. 10 and 11. The measurement results in Fig. 10 clearly show that a tunable bandstop behavior is achieved when the reverse bias applied to the varactor diode is changed from 0 volts to 30 volts while the pair of PIN diodes are switched OFF. The insertion loss above the resonance frequency of the S-SRR is due to the Ohmic loss arising from the finite impedance of the PIN diodes in their OFF state and to the effects of the resonator, which modifies the impedance of the host line. Figure 11 shows that, alternatively, a tunable passband is achieved by tuning the varactor’s bias voltage while the pair of PIN diodes are switched ON by applying a forward

Frequency (GHz)

Fig. 10. Measured transmission coefficients of the reconfigurable and tunable structure of Fig. 9 for different reverse bias voltages applied to the varactor diode while the shuntly connected PIN diodes are in zero bias voltage (i.e. in OFF state).

Frequency (GHz)

Fig. 11. Measured transmission coefficients of the reconfigurable and tunable structure of Fig. 9 for different reverse bias voltages applied to the varactor diode while the shuntly connected PIN diodes are in forward bias (i.e. in ON state).

bias voltage. Thus, the reconfigurability and tunability of the structure are experimentally validated.

III. APPLICATION OF S-SRRS TO UWB ANTENNAS WITH

TUNABLE NOTCH BAND

This section is focused on the application of the S-SRR in the design of UWB monopole antennas with tunable notch bands. To this end, a standard UWB elliptical monopole with a CPW feedline is used as baseline antenna. Next, it is shown that a tunable notch band can be introduced by loading the CPW feedline of the antenna with a pair of varactor-loaded S- SRRs. To maintain a high efficiency for the UWB antenna, the PIN diodes for bandpass/bandstop reconfigurability are not included in this design. In order to demonstrate the scalability of the proposed method, UWB monopole antennas with tunable notch band to reject the interference with the WiMax (3.3–3.6 GHz) and WLAN services (5.15–5.35 GHz or 5.725–5.825 GHz) are presented. In both cases, prototypes of the antennas have been fabricated to validate experimentally the design methodology and simulation results.

A. Baseline UWB Monopole Antenna with CPW Feedline

To exploit the advantages of the S-SRR in terms of compactness, and more importantly, to demonstrate the tunability of the resonator for notch band generation, the baseline UWB antenna needs to be fed with a CPW line.

Fig. 12. (a) Top view of the proposed baseline antenna with CPW feedline. The antenna in composed of an elliptic radiator and a CPW feedline etched on the top side of the substrate, while the copper layer of the back side is entirely removed. (b) The simulated VSWR of the proposed baseline antenna. The VSWR below 2 shows a good impedance matching across the UWB frequency band from 3.1 GHz up to 10.6 GHz.

The considered baseline design is shown in Fig. 12 (a). It is patterned on a 0.257 mm thick Rogers Duroid 5880 substrate with a relative permittivity r = 2:2 and loss tangent tan = 0:0009. The antenna is composed of an elliptic radiator and a CPW feedline, which is realized with a 35 m thick copper layer on the top side of the substrate, while the copper layer on the back side of the substrate is entirely removed. The dimensions of the 50 CPW feedline are w = 2:5 mm and s = 0:2 mm. The dimensions of the substrate and the CPW ground planes are as follows: L1 = 49:4 mm, W = 35 mm, L2 = 23 mm. The dimensions and aspect ratio of the elliptical radiator, as well as the feed gap h between the radiator and the ground planes are optimized to achieve close to a 50 matched input impedance, with a VSWR below 2.0 over the FCC defined UWB spectrum ranging from 3.1 GHz to 10.6 GHz. The optimized dimensions are rmin = 12 mm, rmaj = 16 mm, and h = 0:35 mm.

The simulated VSWR of the proposed baseline antenna is depicted in Fig. 12 (b), showing a good impedance matching across the UWB frequency band.

B. UWB Antennas with Tunable Notch to Reject the Interference with WiMAX or WLAN Systems

Now let us focus on the design of a UWB antenna with a tunable notch for reducing the electromagnetic interference with the WiMAX or WLAN systems. To this end the CPW feedline of the baseline UWB antenna can be equipped with a varactor-loaded S-SRR. The S-SRR could be then tuned to inhibit the propagation of electromagnetic waves along the CPW feedline at the interferer frequency band. Considering the results of Fig. 5, the proposed method has the potential to tune the notch band from 3.1 GHz up to about 5.6 GHz. Thus, ideally, the notch can be used to prevent the electromagnetic interference with either the WiMax or WLAN services.

The feedline of the proposed UWB antenna with a tunable notch is illustrated in Fig. 13, with the antenna being otherwise identical to the baseline antenna. For better visibility, the UWB

antenna is illustrated from the back side, i.e. with the antenna layer (shown in orange color) in the background and the layer of loading S-SRRs (shown in yellow color) in the foreground. As shown in the figure, the feedline is loaded with a pair of S-SRRs in order to achieve a pronounced notch band. Both S-SRRs are loaded with varactor diodes, which are biased through the DC lines connected to the sides of the S-SRRs. The DC bias can be isolated from the RF signal by RF chokes or high impedance resistors. The dimensions of the S-SRRs, which are adjusted to roughly resonate at the WiMax band are as follows: a = 5:3 mm, b = 4:3 mm, d = 1:2 mm, g = 1 mm, c = 1 mm. Both bandwidth and rejection level of the notch band are affected by the distance between the pair of S-SRRs. Thus, to achieve the best performance the space between the two S-SRRs is set to a value e = 1:8 mm optimized through a parametric study. Note that the varactor diodes on the two resonators are oriented in a way so that the asymmetry of the two S-SRRs cancels each other. Further, to suppress the slotline mode, that can be excited by the asymmetric structures, the CPW feedline is also loaded with an air-bridge. The distance between the air-bridge and the end of the feedline is m = 1 mm. It is important to note that apart from some local modification of the slot width of the CPW feedline, which is required to compensate for the capacitive loading of the air-bridges, loading the baseline UWB antenna with the S-SRR does not change the frequency response of the antenna. The fact that the performance of the antenna remains nearly unaffected outside of the tunable band-notch is an important feature of the proposed concept.

The simulated VSWR of the proposed antenna for different values of the varactor capacitance C = 0:05 pF to 2.7 pF are depicted in Fig. 14. The simulation results show a good impedance matching (VSWR below 2) throughout the UWB band except for the narrow notch band close to the resonance frequency of the S-shape SRRs. The simulation results also clearly show that the frequency of the notch band can be tuned in a wide range of 3.1 GHz to 5.6 GHz, where several narrow band interferer services exist, including WiMax and WLAN. Moreover, the simulation results confirm that loading the feedline with the S-SRRs and tuning the frequency of the notch band have negligible effect on the response of the antenna at other frequencies. It is observed that the lower skirt in the VSWR has a much sharper slope compared to the upper skirt. This sharp slope in the lower skirt is due to the presence of a reflection zero (complete transmission) below the resonance frequency of the S-SRR [25].

In order to investigate the effect of the proposed technique on the time domain characteristics of the antenna, a pair of the proposed antenna, which are spaced 20 cm apart, are simulated and the transmission group delay of the setup is plotted in Fig. 15. Variations of the simulated group delay in the unnotched frequency band is small, indicating a linear phase in the far-field region and no sensible pulse distortion at desired UWB frequencies.

In order to validate the simulation results, a prototype of the proposed antenna has been fabricated and measured. Figure 16 depicts photographs of the top and bottom sides of the prototype. For tuning purposes, the S-SRRs are loaded

Fig. 13. The layout of the proposed UWB antenna feed with a tunable notch to reject interferer systems operating in the UWB frequency band, such as WiMax and WLAN services.

Fig. 16. Photographs showing top and bottom views of fabricated prototype of UWB antenna with a tunable notch-band. The figure also shows the zoomed picture of the varactor-loaded resonator and its biasing network. It is worth mentioning that in the prototype three more airbridges have been fabricated for testing purpose. However, the measurement was performed while these airbridges were disconnected

Frequency (GHz)

Fig. 17. Measured VSWRs of the proposed antenna for different values of

the DC bias voltage from Vbias = 0 V to 30 V. The inset shows the full UWB band to illustrate that higher frequencies in the band remain unaffected.

Fig. 14. Simulated VSWR of the proposed antenna for different values of the varactor capacitance C = 0:05 pF to 2.7 pF.

Frequency (GHz)

Fig. 15. Simulated group delay of the proposed antenna (blue solid-line) and the baseline antenna (red dashed-line). The notch band for the loaded antenna is indicated as shaded zone. The inset shows the simulation setup.

with Skyworks [35] SMV1405 varactors. As mentioned earlier, the capacitance range of this diode type (0.63 pF to 2.67 pF) is narrower than the range used in the simulations. The measured VSWRs of the antenna for different values of the varactors’ capacitance (0.63 pF to 2.67 pF) corresponding to different bias voltages from 30 volts down to 0 volts are depicted in Fig. 17. It is noted that, considering the tuning range of the utilized varactor, the measured frequency of the notch and its tuning range are in good agreement with the simulation results of Fig. 14. This tuning range covering the frequency band 3.1 GHz up to about 3.7 GHz is sufficient to reject the WiMax interference. It is also clear that the limited tuning range is due to the technical limitation of the capacitance range for the utilized varactor, and is not a limitation of the proposed method. As was demonstrated above in the simulations, a varactor capacitance range of 0.05 pF to 2.7 pF would allow covering a notch tuning range extending from 3.1 GHz to 5.6 GHz, i.e. sufficient to reject interference from systems operating in this frequency band, including WiMax and WLAN services. The measured VSWR of the prototype for the whole UWB band is depicted as an inset

in Fig. 17, confirming that tuning the frequency of the notch has no adverse effect on the VSWR of the antenna at other frequencies.

In order to demonstrate the scalability of the proposed method, while using the same varactor type, a second UWB monopole antenna with a tunable notch band to reject the interference with WLAN services has been designed. The layout of the proposed UWB antenna is identical to the previous antenna illustrated in Fig. 13. The substrate and dimensions of the baseline antenna remain unchanged, while the dimensions of the S-SRRs are scaled to a = 3:5 mm, b = 2:5 mm, d = 1:3 mm, g = 0:7 mm, and c = 0:2 mm. The spacing of the S-SRRs and the air-bridge are e = 1:8 mm, and m = 1 mm.

The simulated VSWR of the second antenna for different values of the varactor capacitance C = 0:5 pF to 2.7 pF are depicted in Fig. 18. The simulation results show a good impedance matching (with VSWR below 2) in the UWB band, with the exception of a tunable narrow band close to the resonance frequency of the S-shape SRRs. This notch can be used to reject the interference with lower WLAN (5.15–5.35 GHz) or higher WLAN (5.725–5.825 GHz) services, without affecting the response of the antenna at other frequencies.

In order to validate the simulation results, a prototype

(c)

Fig. 20. Simulated (solid lines) and measured (dashed lines) far-field radiation patterns of the proposed band-notched UWB antenna at (a) 4 GHz, (b) 7 GHz, and (c) 10 GHz. The left column shows the H-plane (xy-plane) patterns, whereas the E-plane (xz-plane) radiation patterns are depicted on the right. The blue lines represent the co-polarization, while and the red lines show the cross-polarization.

of this second antenna has been fabricated and measured. The measured VSWRs of the antenna for different values of the SMV1405 varactors’ capacitance (0.63 pF to 2.67 pF) corresponding to different bias voltages from 30 volts down to 0 volts are depicted in Fig. 19, showing a good agreement with the simulation results of Fig. 18.

Figure 20 depicts the simulated (solid-line) and measured (dashed-line) normalized co-polarized (blue) and crosspolarized (red) far-field radiation patterns of the proposed UWB antenna at 4, 7, and 10 GHz. For the sake of brevity, results are only shown for the first prototype, i.e. with notch band in the WiMAX band. In each case the figure on the left shows the radiation pattern in the H-plane, whereas the E-plane radiation pattern is depicted on the right. As expected, the figures demonstrate that the proposed antenna has a reasonably omnidirectional pattern in the H-plane, while the radiation pattern in the E-plane exhibits a dipole behavior typical of UWB antennas, with some degradation at higher frequen-

Fig. 21. Measured and simulated maximum realized gains, as well as the simulated total efficiency of the proposed antenna for the DC bias voltage Vbias = 0 volts, corresponding to a notch band at f = 3:2 GHz.

cies. The simulated patterns with and without the tunable S-SRRs are not significantly changed from the ones of the baseline antenna (not shown for brevity). The measured and simulated maximum realized gains, as well as the simulated total efficiency of the proposed antenna with a notch-band tuned at f = 3:2 GHz (corresponding to the DC bias voltage Vbias = 0 volts) are presented in Fig. 21. The figure shows that the antenna has an acceptable realized gain and high efficiency over the UWB frequency band, with the exception of a strong notch at the resonance frequency of the S-SRR.

A comparison between the proposed antenna with several UWB antennas in [5]–[10], [12] is presented in Table II. The comparison highlights some features of the proposed method:

1)In terms of physical dimensions, the proposed antenna is a planar structure, that is slightly larger than [8], [9] (among the tunable antennas) but smaller than those in [5]–[7]. However, it is important to note that the larger physical dimensions of the proposed antenna are not due to the loading resonators. Also we note that the very compact size of the antenna in [12] is mostly due to utilizing a fabrication technology with a feature size smaller than 1 m. In fact, the CPW fed baseline antenna from [9] or [12] could be used to reach the same level of miniaturization as in those antennas using the proposed S-SRRs.

2)While the miniaturization of the proposed UWB antenna is limited by the baseline antenna, the compact size of the proposed resonator has allowed us to use a pair of S- SRRs (rather than a single S-SRR), resulting in a VSWR consistently higher than 8. This translates into more than 10 dB decrease in the realized gain within the notch band.

3)A notch band with a large tuning range is achieved. This large tuning range arises not only from tuning of a variable capacitor, but also from a continuous change of topology of the resonator, from a pair of SRRs to a single S-SRR.

4)Finally, a comparison between the radiation patterns of the UWB antennas with tunable notch band in [5]–[9] reveals that, generally the far-field radiation patterns of the UWB antennas with a tunable slot in the radiator are more distorted than the patterns of the antennas with a filtering mechanism in the feedline (including the proposed antenna).

IV. CONCLUSION

In summary, the application of S-SRRs excited by contradirectional magnetic fluxes of a CPW has been demonstrated for reconfigurable and tunable operation. Specifically, by loading an S-SRR with a PIN diode the resonance frequency of the resonator can be switched to that of a pair of SRRs. On the same bases, a tunable range of almost one octave can be achieved if the S-SRR is loaded in its center with a varactor diode. Furthermore, it has been demonstrated that a reconfigurable (bandstop or bandpass) device can be achieved if a pair of shuntly connected PIN diodes are introduced across the slots of the host CPW. This feature, in conjunction with the tunability of a varactor-loaded S-SRR, has been used to achieve a reconfigurable and tunable structure. Finally, in order to demonstrate the potential application of the proposed structure, a UWB antenna design with tunable notch band for WiMax or WLAN services interference rejection has been demonstrated. The design methodology has been validated through electromagnetic simulations and measurements of fabricated prototypes.

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