диафрагмированные волноводные фильтры / ddbbaea3-48bc-47e0-a2a7-7a666af75369

ABSTRACT—This letter presents a substrate integrated waveguide (SIW) based compact, dual mode self-diplexing antenna. A single square cavity with a rectangular ring slot is used for both dual mode excitation and radiation. A prototype diplexing antenna is fabricated that radiates linear, orthogonal polarization at 9.5 GHz and 10.5 GHz. The measured peak gains at the two frequencies are 5.75 dBi and 5.95 dBi, respectively. Intermodal coupling between the modes provides two transmission zeros, which are utilized to achieve port isolation better than 29 dB. The proposed technique can be used for closely spaced radiating frequencies. As no external filter is used, the structure is compact. Further, because of the slot loading effect, the antenna is smaller than a recently published antenna claimed as the most compact antenna. Its working principle is discussed in detail. Finally, design guidelines are suggested.

INDEX TERMS— Diplexer, dual-band antenna, dual mode resonator, substrate integrated waveguide (SIW).

I. INTRODUCTION

M odern communication systems demand compact, low profile, dual, and multiband antennas. On the other hand, the SIW offers compact circuit size and lower loss [2]. These advantages make SIW based antennas popular. In [3], dual-band SIW antennas with triangular ring slot are presented. The triangular slot on top plane supports slot and patch modes which radiate at 9.4 GHz and 16 GHz, respectively. In [4], a dual-band circularly polarized antenna that radiates through a circular slot at 10 GHz and 12.5 GHz has been reported. A dual band MIMO antenna is presented in [5]. The first resonance is at 2.4 GHz. The 2nd and 3rd resonances are combined for wider bandwidth at 5 GHz. Dual mode circularly polarized SIW antennas using cross slot are reported in [6]-[7]. In [8], dumbbell slots on top of a SIW cavity are used for dual frequency operation at 9.4 GHz and 13.62 GHz. In [9], conventional and modified TE120 cavity modes are used for the dual-band, linear and circular polarization at 13.4 GHz and 17.9 GHz. However, all the dualband antennas in [3], [5], and [9] offer poor port isolation. Port isolation of dual frequency antenna and mono pulse antenna array are improved by using dual mode and third order Chebyshev filters, respectively [10]-[11]. In [12], dual polarized microstrip antenna array is reported over 2.7 - 2.9 GHz for phased array radar applications. The aperture coupled

feed for horizontal and differential probe feed for vertical

The authors are with the Electronics and Electrical Communication Engineering Department, Indian Institute of Technology, Kharagpur, West Bengal, India, 721302 (e-mail: arani.ali.khan@gmail.com, mkmandal@ieee.org).

Fig. 1. Layout of the proposed antenna.

polarization improves the port isolation to 45 dB. In [13], a dual-polarized 1×2 patch antenna array is presented over 2.5- 2.7 GHz. A pair of cavities below the radiating elements and an artificial periodic structure improve port isolation to 45 dB.

On the other hand, self-diplexing antennas offer good isolation between its two ports while operating at two different frequencies. A dual band antenna at 2.45 GHz and 5.5 GHz with double T-stub loaded aperture is presented in [14]. A self-diplexing antenna is proposed for GSM and Digital Cellular System bands [15]. In [16], two ports of a selfdiplexing antenna are directly connected to LNA and power amplifier at 5.4 GHz and 5.5 GHz, respectively, without any diplexer. In [17], two HMSIW resonators are placed together with a common slot for radiations at 8.97 GHz and 11.3 GHz. Isolation between the two ports is around 20 dB. However, the designs of [14]-[17] are not suitable for closely spaced channels, typically within 5% to 10% of the average frequency, frequently seen in satellite applications. Microstrip based duplexing antennas with extra band pass filters (BPF) are reported in [18]-[19]. In [20], a square SIW cavity with two orthogonal ports achieves dual polarizations. But, both the ports excite the antenna at the same frequency of 2.4 GHz. Recently, in [21], a self-diplexing SIW antenna is presented using two slots for radiations at 8.26 GHz and 10.46 GHz. However, isolation between the ports decreases with deceasing frequency separation.

Here, a compact, dual mode, self-diplexing SIW antenna is presented using a single SIW square cavity with orthogonal feed lines. The intermodal coupling between two modes, popular in filter design, is used to create transmission zeros (TZ). The TZs are then utilized to greatly improve the port isolation. In contrast to conventional designs, the proposed technique is best suitable when two narrow communication channels are closely spaced. The tuning of resonant frequencies, isolation, input matching and radiation characteristics are discussed in detail.

II. DUAL MODE SIW CAVITY ANTENNA

Fig. 1 shows the proposed SIW antenna with a rectangular slot etched on top. Port-1 and Port-2 excite TE210 and TE120 resonant modes, respectively. A 1.58 mm thick RT/Duroid 5880 substrate (εr = 2.2, tanδ = 0.0009) and the Ansoft’s HFSS full wave simulator are used for all the studies.

A. Dual Mode SIW cavity and the radiator

At first, a square cavity of dimensions W = L = 20.6 mm, via diameter d = 1.0 mm, and pitch p = 2.0 mm is designed for f210 = f120 = 11 GHz. The cavity is excited by two 50 Ω microstrip lines of width wm = 4.58 mm. Initially, the feed offsets x1 = x2 = 0 mm. The feed gaps xf1 and xf2 control the external Q-factors and hence the input matching at the two ports. Their effect on resonant frequencies is insignificant. Without any perturbation, the modes are orthogonal to each other and hence they are isolated. The |S|-parameters of this cavity are shown in Fig. 2(a). The field plots in Fig. 2(b) and

(c) show that an electric wall bisects the unused port symmetrically which results a port isolation of at least 25 dB.

Next, as shown in Fig. 1, a square ring slot of dimension a = b = 8.0 mm and ws = 1.0 mm is placed symmetrically about the central axis. The offsets x1 and x2 are kept zero as before. The slot perturbation introduces mode coupling and hence the port isolation is lost. As the structure is symmetric with respect to the ports, TE210 and TE120 modes have the same resonant frequency of 10 GHz. Slot loading decreases resonant frequency. Fig. 3(a) shows the |S|-parameters for different dimensions of the square slot. The plots show that always |S11|

=|S22|. Due to the symmetry of the structure, excitation at two ports still results f210 = f120. Fig. 3(b) and 3(c) show vector H- fields for port-1 and port-2 excitations, respectively, for a = b

=8.0 mm. It can be observed that the electric wall still bisects the unused ports though asymmetrically that results in small transmission between the ports and the port isolation decreases to 10 dB. However, because of signal cancellation due to the different signal paths created by TE210/TE120 and lower order TE110 modes inside the cavity, a TZ is created at a frequency higher than the resonant frequency that results in a dip in |S21|.

Next, the square slot is changed to a rectangle of

dimensions ws = 1.0 mm, a = 6.0, b = 8.0, so that the outer peripheral length is approximately λg, where λg is the guided wavelength inside the SIW at the average resonant frequency. For proper input matching, the feed gaps are xf1 = 7.21 mm and xf2 = 8.12 mm. Fig. 4(a) shows the |S|-parameters. Rectangular slot makes the structure asymmetric about the two

ports and results f210 = 10.45 GHz and f120 = 9.62 GHz. Fig. 4(b) and (c) show the vector H-field distribution at these two frequencies. The field distributions show even and odd symmetries about the central plane passing through the port of excitation and the unused port, respectively. It is interesting to observe that the rectangular slot improves port isolation due to mode separation and two TZs at 10.0 GHz and 10.75 GHz. The zeros TZ1 and TZ2 are created by the intermodal coupling between TE210/TE120 and the fundamental TE110 modes [22]. The TZs can be tuned by the offsets x1 and x2 for further isolation improvement. The fundamental TE110 resonant

(a) (b) (c)

Fig. 2. (a) |S|-parameters of a square cavity without any perturbation. H-field isolines for (b) port-1 and (c) port-2 excitations (f210 = f120 = 11.0 GHz).

(b) (c)

Fig. 3. (a) |S|-parameters of the square cavity with a square ring slot with arm length 8.0 mm. H-field isolines for (b) port-1 and (c) port-2 excitations.

(a)

excitation (f210 = 10.45 GHz) and (c) port-2 excitations (f120 = 9.65 GHz).

frequency of this slot loaded cavity is f110 = 7.06 GHz. Surface current and vector E-field distributions at slot at f210 and f120 are shown in Fig. 5. It is evident from the plots that at both the frequencies the ring slot behaves as one wavelength loop that provides linear polarization having far-field E-field parallel to the cavity wall containing the port of excitation. However, because of the change in even and odd mode excitation planes, position of current maxima changes with port number. Thus,

> A. A. Khan et. al. <

(c) (d)

Fig. 5. Surface current distribution across the slot with a = 6.0 mm and b = 8.0 mm for (a) port-1 excitation ( f210 = 10.45 GHz), (b) port-2 excitation (f120 = 9.65 GHz) and vector E-field distributions at slot arms for (c) port-1 excitation ( f210 = 10.45 GHz), (d) port-2 excitation (f120 = 9.65 GHz).

radiated electric fields at the two frequencies are orthogonal. Dimensions of the patch inside the slot along a and b are 0.22λg and 0.28λg at the mentioned frequencies, which are considerably lower than the required value of 0.5λg for radiation from a patch. Fig. 5(a) and (b) also show that there is no current distribution across the patch. Fig. 5(c) and (d) show that the required E-field across the slot arms for radiation at two frequencies are created by the slot loaded cavity modes. Therefore, the radiation frequency is determined by both the slot and the cavity dimensions.

B. Tuning of resonant frequencies

Fig. 6. Variation of f210 and f120 with (a) a for b = 8.0 mm, and (b) b for a = 6.0 mm. (W = L = 20.6 mm, xf1 = 7.16 mm, xf2 = 7.76 mm and x1 = x2 = 0.0 mm).

Fig. 6 shows that f210 and f120 can be tuned by a and b, respectively. The f210 changes by 0.9 GHz when a is changed from 4.5 mm to 7.5 mm. At the same time, change in f120 is 0.16 GHz. This is due to the odd symmetry of E-field distribution that places the electric wall parallel to the a arms. Therefore, for f210 excitation, the capacitance due to the patch is mostly affected by the dimension a. Similar explanation holds for port 2 excitation. By a proper choice of a and b, the frequency ratio f210/f120 can be tuned between 1.8% to 15.2%. The dimensions a and b are set to 6.5 mm and 8.9 mm, respectively, for desired f210 = 10.5 GHz and f120 = 9.5 GHz.

C. Improvement of isolation

Fig. 7(a) shows that ws has negligible effect on isolation. Keeping the outer perimeter of the slot fixed, variation in ws changes the area of the patch inside the slot which changes the

3

capacitive loading of the cavity hence the resonant frequencies. The effects of ws on radiation characteristics are tabulated in Table-I. The slot width ws is optimized for maximum gain and minimum cross-pol level. The optimized ws = 1.0 mm also sets f210 = 10.5 GHz and f120 = 9.5 GHz.

Next, the feeding offsets x1 and x2 are optimized for best isolation keeping other dimensions fixed. Two TZs at 10 GHz and 10.75 GHz shown in Fig. 4(a), are shifted to 9.4 GHz and 10.65 GHz, respectively, as shown in Fig. 7(a), due to the tuning of resonant frequencies by a and b. As the TZs are close to the resonances, a minimum port isolation of 24 dB has been obtained. The x1 and x2 control the strength of intermodal coupling between the higher order TE210/TE120 and the lower order TE110 mode. The coupling strength of TE110 mode can be increased by keeping the two ports close to each other by tuning x1 and x2. As a result the TZs shift to lower frequencies [22]. The offsets x1 and x2 are considered positive when they are in positive Y and X directions, respectively. To bring the two ports close to each other, only positive x1 and negative x2 values are considered. Fig. 7(b) shows the simulated |S|- parameters for different x1 and x2. It shows that the port isolation improves to 28 dB for x1 = 1.0 mm and x2 = -1.0 mm. The effect of x1 and x2 on f210 and f120 is insignificant. However, the offsets change the positions of magnetic wall for TE110 mode and its effect on f110 is more prominent. The frequency f110 changes from 7.06 GHz for zero offsets to 7.50 GHz for x1 = 1.0 mm and x2 = -1.0 mm. Since f110 comes close to the main resonances, the coupling strength of the lower order mode increases which shifts the TZs to lower frequencies and improves the port isolation.

TABLE I

EFFECT OF SLOT WIDTH ON ANTENNA PARAMETERS

The above studies suggest the following design steps.

•Calculate the dual mode square cavity dimensions W and L at (f210 + f120)/2.

•Place the rectangular ring slot at the centre of the cavity

as shown in Fig. 1. Initial values for the slot are 2(a + b) = λg at average frequency and ws is approximately λg/20 at the higher frequency; λg is the guided wavelength inside SIW.

•Tune a and b for the required f210 and f120.

•Optimize slot width ws for best radiation characteristics.

•Optimize x1, x2 for best isolation.

•Tune xf1 and xf2 for input matching.

•Finally, the cavity dimensions and/or slot arm lengths can be fine tuned to adjust the resonant frequencies.

III. FABRICATION AND MEASUREMENT

Following the design steps, a self-diplexing antenna with

f210 = 10.5 GHz and f120 = 9.5 GHz is designed. The optimized dimensions are a = 6.0 mm, b = 8.64 mm, 2d = p = 2.0 mm, L

= W = 20.72 mm, x1 = 0.9 mm, x2 = -0.9 mm, xf1 = 6.78 mm, xf2 = 7.85 mm, wm = 4.58 mm and ws = 1.0 mm. The prototype is fabricated on a 1.58 mm thick RT/Duroid 5880 substrate. Fig. 8(a) and 8(b) show its photograph and the simulated and measured |S|-parameters, respectively. Measured isolation between two ports is at least 29 dB at two resonances. The 10 dB impedance matching bandwidths (BW) are 1.32% and 1.46% at f210 and f120, respectively. This narrowband makes the antenna suitable for use in a transceiver system with closely spaced narrowband channels. Fig. 9 shows the simulated and measured radiation patterns of the antenna. At the two frequencies, the antenna radiates linear polarization in orthogonal directions. The measured peak gains at f210 in E- plane and H-planes are 5.95 dBi and 5.75 dBi, respectively. The cross-pol levels at the direction of maximum gains are down by 24 dB and 30 dB in E-plane and H-planes, respectively. For f120, measured peak gains on E-plane and H- plane are 5.55 dBi and 5.75 dBi, respectively, with the corresponding cross-pol suppressions as 28 dB and 32 dB. Due to the slight asymmetry introduced by the offsets x1 and x2, the E-plane patterns are slightly tilted in opposite direction.

IV. CONCLUSIONS

The proposed antenna is based on dual-mode resonances of a SIW square cavity. The dual resonances and proper placement of TZs help to achieve high port isolation without the use of extra BPFs. The BW and isolation of the proposed design can easily be further improved with the use of extra cavities [18]-[19]. Finally, a comparison with other work in Table II show that the proposed design offers closest band separation, best isolation, gain, and cross-pol level without any

(a) (b)

Fig. 8. (a) Photograph of the fabricated structure and (b) |S|-parameters of the optimized structure.

Fig. 9. Simulated and measured radiation pattern of the optimized structure at f210 = 10.5 GHz on (a) φ = 00 (E-plane), (b) φ = 900 (H-plane) and at f120 = 9.5 GHz on (c) φ = 00 (H-plane), (d) φ = 900 (E-plane).

extra BPFs in SIW technology. This proposed SIW antenna is also more compact in size (λg × λg) than the most compact (1.04λg × 1.35λg) self-diplexing SIW antenna of [19].

TABLE II

COMPARISON WITH OTHER REPORTED WORKS

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