диафрагмированные волноводные фильтры / 9f9c9e94-9c56-40c4-97a8-950c43198e70

Abstract — A W-band Schottky diode based frequency tripler which uses waveguide resonator filters for low loss impedance matching is presented in this paper. Impedance matching of the diodes is achieved by scaling the external quality factors and adjusting the resonant frequencies of the filter cavities. This removes most of the matching structures from the high loss microstrip circuit to the lower loss waveguide resonators. Here the output frequency of the tripler is set to be 90 GHz with a 10% bandwidth. The simulation shows a conversion loss of 13-13.8 dB in the pass-band with an input power of 13–20 dBm. The measured conversion loss over the pass-band is 13.6 –15.8 dB for 17 dBm input power and better than 14 dB at 90 GHz for 13-20 dBm input power.

Keywords —coupling matrix, frequency multiplier, impedance matching, waveguide resonators.

I.INTRODUCTION

Filters and impedance matching networks are frequently used microwave elements. In conventional designs, filters and matching networks are usually designed separately and then arranged in series to fulfil their functions individually. However, the concepts of impedance matching and filtering are not intrinsically independent to each other and components with integrated filtering and impedance matching functions are also possible. Examples are impedance transformers with low-pass or band-pass responses are given in [1]-[2]. More examples are filters with complex load impedances, which are designed to directly match a load with complex impedance while maintaining the intrinsic filtering function. These matching filters can be described by lumped LC model as discussed in [3] or coupling matrices as shown in [4]-[6]. The latter can be can be used to design filters using any kind of resonator, bringing great flexibility to the technology used [7]. This method is already adopted in some power amplifier designs (working at a few GHz) presented in [8]-[9], where the input and output impedances of the active components (e.g. transistor) are directly matched using bandpass filters. The use of high quality factor (Qu) resonators in filters to directly match the devices eliminates the need of an extra matching network. This leads to a reduced circuit complexity, circuit size and a lower loss.

In this paper, we extend the idea to the design of frequency conversion devices, with an example of a Schottky diode

(a)

(b)

Fig. 1. Topology of the tripler with waveguide resonator matching. The input and output of the diodes are coupled to the 3rd and the 4th resonators.

(a) Physical structure of the tripler, including the input BPF, the diodes and the output BPF. (b) Diagrammatic view of the resonators and the couplings. The diode chip used here is UMS DBES105a, it contains two series connected anodes with Cj0=9.5 fF.

frequency multiplier, operating at millimetre-wave frequency, where the loss due to conversional planar matching circuits can become significant. Here, a W-band tripler is given as an example, as shown in Fig. 1. In this design, the input and output of the Schottky diodes are coupled to the 3rd and the 4th resonators, via two E-plane probes on the microstrip. Impedance matching is realized within the filters by scaling the external quality factors and adjusting the resonance frequency of the filter cavities[4]-[6], [8]-[9].

II. DESIGN OF THE TRIPLER

The tripler is designed to have a 10% FBW centred at 90 GHz (at the output port) with a nominal input power of 17 dBm. As shown in Fig. 1 (a), the tripler is comprised of three parts: The input 3rd order filter, the diodes and the output 2nd order filter. The diodes working at varistor mode are matched by using these two filters. Fig. 2 shows the input and output impedances of the diode (the extraction process for these

(a)

(b)

Fig. 2. Input and output impedance of the diodes. (a) Input impedance.

(b) Output impedance. The impedances are obtained under 17dBm input power.

(a)

(b)

Fig. 3. Input and output filters. (a) input filer. (b) output filter. The dimensions in millimetre are : L1=2.55, L2=8.98, L3=1.93, L4=9.51, L5=2.33, L6=5.53, L7=2.58, L8=2.23, L9=0.96, L10=3, L11=0.94, L12=2.41, L13=0.83, L14=0.48.

impedances will be discussed later). Different to conventional waveguide filters, for which the load impedances are real numbers, here the diode has complex input and output impedances. As shown in Fig. 1 (a), the input and output impedances of the diodes are defined as Zin and Zout at the positions marked by dashed lines. They are valid and well

defined on the microstrip line but when transformed to the waveguides via the E-plane probes, the impedance presented to the waveguide resonator is unknown. This is why a coupling matrix approach is superior to impedance considerations.

To address the issue, recall the general procedure of designing a microwave filter using coupling matrix [7]: The matrix can be found from the filter specification and the information contained in the matrix is the coupling coefficients between resonators (Mij) and the external quality factors (Qe) between resonators and the source/load. The relationship between Qe and the physical dimensions of the filter can be found using full-wave simulators, such as CST [10]. This is generally done by simulating a single resonator and then finding the required Qe from the S21 response. Taking the input filter as an example, the Qe of the filter is controlled by the length of the probe (see Fig.3 (a), marked as L7) and the resonant frequency is controlled by the length of the resonator (Fig.3 (a), marked as L6). The length of the probe and the resonator can be tuned in CST to get the wanted Qe. Hence, the dimensions of the resonator can be altered to get the correct Qe under the presence of the complex impedance. In other words, the complex impedance can be compensated directly during the full-wave simulation and the consequences are the altered resonator dimensions. The design details of the tripler will be presented as follows

A. Extraction of the input and output impedances:

The input and output impedance of the tripler is directly matched by the filters, hence for the design, the impedances of the diodes must be extracted first (at the 1st and the 3rd harmonic). As shown in Fig.1 (b), two Schottky diode chips (UMS DBES105A, working at varistor mode) are placed in anti-parallel configuration, each with two anodes in series. The three-dimensional model of the diode chip together with its waveguide housing is modelled in CST. The S-parameter file of the structure is extracted and imported into circuit simulation software such as ADS [11], where the non-linear Schottky junction can be modelled. Harmonic balance is used to extract the input and output impedances of the diodes [12]. The extracted impedances are shown in Fig.2. They are used as the load impedances for the filters described in the next section.

B. Design of the input filter.

The input filter is shown in Fig. 3 (a). The filter is a 3rd order filter with Chebyshev response. The return loss is chosen as - 20 dB and the bandwidth is defined as 10% centred at 30 GHz. The coupling matrix is used to design of the filter and the matrix is [7]

With Qe1 = Qe3 = 8.51. These coupling coefficients can be used by standard filter techniques [7] to find the dimensions marked as L1 to L8 in Fig. 3 (a). Note that the input impedance of the diodes (Zin) is assigned to the microstrip port of the filter

(a)

(b)

Fig. 4. Responses of the matching filters. (a) input filter. (b) output filter.

during the full-wave simulations (ZL=Zin), as shown in Fig.3 (a). The optimized filter response can be found in Fig. 4 (a).

C. Design of the output filter.

The output filter is shown in Fig. 3 (b). This 2nd order filter is designed to have a Chebyshev response with a passband return loss of 20 dB, a fractional bandwidth of 10%, and a centre frequency of 90 GHz. Similar to the input filter design, the coupling matrix can be calculated as

With Qe1 = Qe3 = 6.64. Again, this can be used to estimate the dimensions of the filter. Then the filter is optimised in a fullwave simulator, with the load port impedance (ZL) equal to that of the output impedance of the diode (Zout). The optimized filter response is shown in Fig.4 (b).

D. Simulated performance of the tripler.

The simulated performance of the tripler together with the matching filters can be predicted by combining the S-parameter files of the filters (obtained using CST) with the non-linear Schottky diode model in ADS. Fig. 7 shows the simulation results of the whole device. The conversion loss is 12.5-13 dB, when the input power is 17 dBm. The simulated S11 is better

(c)

Fig. 5. Photos of the fabricated device. (a).The metal box. (b).The bottom half of the device. (c).Enlarged view of the substrate.

Fig.6. The measurement setup

than -20 dB and has three distinguishable reflection zeros, across the passband. The appearance of the reflection zeros demonstrates that the input filter provides excellent, accurate matching to the diode.

III. MEASUREMENT RESULTS

The waveguide structure is machined from aluminium, and the substrate for the microstrip circuit is Rogers Duroid 5880 with a thickness of 0.127 mm. Fig. 5 shows some photographs of the components.

Measurements are performed for input frequencies of 26-34 GHz, at an input power level of 50 mW (17 dBm). To measure the reflection at the input port (i.e. S11), a bi-directional coupler is placed between the output of the power amplifier and the input of the tripler, as shown in Fig. 5. The measured output power as well as the S11 are exhibited in Fig. 6. The measured S11 is better than -10 dB while the conversion loss is 13.6-15.8 dB (4.3%-2.6% efficiency), across the passband. The measured conversion loss is approximately 0.5-2dB higher than expected from simulations. This can be attribute to several reasons:

(i). The 0.5 dB more loss can be attribute to the higher RF resistance of the anodes than it’s DC value we use in the simulation. (ii). The higher losses at some frequencies (up to 2 dB more than simulation) may came from poorer matching from the output filter (which can hardly be measured in this work). Fig. 7 shows the output power of the tripler as a function

Fig.7. Simulated and measured results of the tripler with 17dBm input power

Fig.8. Output power versus input power for 3 frequency points.

of input power levels, for three frequency points, i.e. 29 GHz, 30 GHz and 31 GHz. It can be observed from Fig. 7 that the tripler is capable of offering low conversion loss at a wide range of input power levels, for instance at 30GHz the conversion loss remains better than 14dB with an input power up to 20 dBm.

CONCLUSIONS

A passive W-band frequency tripler directly impedance matched by waveguide filters has been presented in this paper. The coupling matrix is used to design the filters, and the effects from the complex impedance are directly compensated in the full-wave simulation. The simulation shows a conversion loss from 13-13.8dB with 10% of bandwidth while the measured conversion loss is from 13.6-15.8dB within the 10% of bandwidth. This paper demonstrates that filters can be used to directly match some non-linear semiconductor devices while retaining their functional filtering performance. Due to the use of the coupling matrix, the physical structure of the filters can be easily realized by low loss waveguide resonators and this gives a potential to achieve better performance than conventional devices with the filtering and impedance matching networks realized on relatively high loss microstrip/coplanar

circuits, especially in high frequency applications up to submillimeter wave or even terahertz range.

ACKNOWLEDGMENT

The authors would like to thank Prof. Xiaochuan Zhang and Prof. Yuliang Dong from UESTC for their help with the fabrication and Dr. Byron Alderman from Rutherford Appleton Laboratory (RAL) for the help with measurements.

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