диафрагмированные волноводные фильтры / 1c31a32d-f1bc-4cc5-bc05-54c125ee5f14

Abstract—A differential substrate integrated filtering dielectric resonator antenna (DRA) fed by TE20-mode in substrate integrated waveguide (SIW) for 5G millimeter wave (mm-Wave) systems is proposed. The dielectric resonator (DR) acts as a radiator as well as the last resonator of a bandpass filter simultaneously to realize a filtering antenna. The DR operates at the typical differential TE111 mode and is excited by TE20 mode of SIW. The differential feeding scheme provides the antenna with an improvement on radiation pattern symmetry and low cross-polarization level. Besides, this antenna can be manufactured using standard printed circuit board (PCB) technology which minimizes assembly errors. For demonstration, a prototype based on the proposed design theory is designed and obtains an impedance bandwidth of 10.3% from 24.7 GHz to 27.4 GHz, an average gain of 6.7 dBi and a cross-polarization level lower than -35 dB in E-plane and - 20 dB in H-plan.

Keywords—differential antenna, substrate integrated technology, filtering antenna, dielectric resonator antenna (DRA), 5G, millimeter wave (mm-Wave)

I. INTRODUCTION

Recently, the International Telecommunication Union has announced the millimeter wave (mm-Wave) frequencies for modern 5G communication systems [1]. As systems shifting to mm-Wave bands, a higher signal-to-noise ratio has always been desirable for a high date transmission rate and low latency. In this aspect, a differential circuit is attractive to eliminate the common mode interference and enhance the signal transparency [2]-[3]. Meanwhile, the short wavelength in mm-Wave also causes large conductor and surface wave loss which decreases the efficiency of a metallic antenna. In contrast, dielectric resonator antenna (DRA) will be a good candidate to reduce the loss [4].

To be compatible with a high device density in mmWave wireless system, filter and antenna are often integrated into a single module for the size and insertion loss reduction

This work was supported in part by the National Natural Science Foundation of China under Grant 61501265 and 61601250, the Natural Science Foundation of Jiangsu Province under Grant BK20161281, the Postgraduate Research & Practice Innovation Program of Jiangsu Province under Grant KYCX18_2425, and the Nantong University-Nantong Joint Research Center for Intelligent Information Technology.

Fig. 1. Cross-section view of the proposed filtering DRA.

Fig. 2. Side view and layer definition.

[5]. There are mainly three methods to realize a filtering antenna design. The first one is cascading a filter and an antenna which needs a similar impedance at the interface between them [6]. In the second method, the filter part is integrated with the antenna and embedded into the antenna part. The gain response of the antenna is similar to or same as a bandpass filter. Obviously, it is an alternative way to reduce the system size and loss [7]. In the last method, the antenna simultaneously functions as the radiator as well as the last resonator of a bandpass filter. This kind of integration design achieves the maximum reduction of insertion loss and is applicable for 5G mm-Wave

Fig. 3. Dimensions of the DR and metallic ground planes.

communication systems. The input return loss and antenna gain are almost the same as the reference filter [8]. Comparing to the first two methods, it turns out to be the most efficient way in system size and insertion loss reduction.

In this letter, a substrate integrated filtering DRA differentially fed by TE20-mode in substrate integrated waveguide (SIW) is proposed. The filtering antenna obtains the filtering function by setting the radiator as the last resonator of a bandpass filter. Both of dielectric resonator (DR) and SIW cavities are designed to be manufactured using the standard printed circuit board (PCB) technology which minimizes the assembly errors. The intrinsic differential field distribution of TE20 mode in SIW cavities are used to couple each other and excite the differential TE111 mode in the DR from a synthetic three-order filter. The antenna obtains a low cross-polarization level, a symmetrical radiation pattern and a good frequency-selectivity. Simulated results show that the designed prototype achieves an impedance of 10.3 % from 24.7 GHz to 27.4 GHz which is corresponding with the 5G mm-Wave band in 28 GHz, an average gain of 6.7 dBi and a cross-polarization lower than - 30 dB in E-plane and -20 dB in H-plane.

II. ANTENNA CONFIGURATION

The cross section view of the three-order filtering DRA is shown in Fig. 1. Fig. 2 shows the side view and layer definition of the proposed antenna. It consists of one DR and two coupled SIW horizontally cavities. Three layers of substrates with prepreg inserted between them are employed. In the upper part of the configuration, the DR operates at its fundamental TE111 mode and acts as the radiator as well as the last resonator of a passband filter. Metallic vias together with ground planes form two SIW cavities in the lower two substrates. The two SIW cavities operate at TE20 mode and are coupled with each other by a pair of slots etched on the Ground planes 2 and 3 which are separated by a prepreg layer. On the bottom of the structure, microstrip line is used to connect with the input/output port. A typical transition structure is used between the SIW and microstrip line. DRA is contributed by a Rogers 3010 substrate with εr=10.2, tanδ=0.0023 and thickness of 25 mil. Substrate 1 and Substrate 2 are Rogers 5880 with εr=2.2, tanδ=0.0009 and thickness of 10 mil. Layers of prepreg with εr=2.75 and thickness of 4.4 mil are inserted between ground plane 2 and ground plane 3, DRA and ground plane 4. Dimensions of the DR and ground planes are shown in Fig. 3 and all in millimeter.

(b)

Fig. 4. Simulated reflection coefficients and gains of the proposed filtering DRA for different length of slot (ls1). (a) Reflection coefficients. (b) Antenna gains.

(b)

Fig. 5. Simulated reflection coefficients and gains of the proposed filtering DRA for different length of slot (ls2). (a) Reflection coefficients. (b) Antenna gains.

I. PARAMETRIC STUDY

To further demonstrate the influence of the dimensions of slots on the boresight gain and reflection coefficient in detail, a parametric study is carried out by using the simulation software Ansys HFSS. The length of ls1 in Ground plane 4 and ls2 in Ground plane 2 and 3 are the included dimensions. It is noted that when one of the dimensions is changed, the other keep fixed.

Fig. 4 shows the simulated boresight gains and reflection coefficients with different length of ls1 and ls2=2.6mm. In Fig. 4 (a), it is seen that there are three resonant modes in the passband. The first and second resonances are very sensitive to the length of ls1, and they are verified to be caused by the SIW cavities. The change of coupling between the top SIW cavity and DR shows a large deviation to reflection

Fig. 6. Simulated reflection coefficient and gain of the proposed filtering DRA.

coefficient. The third resonant mode which is sensitive to the

DR size is proved to be caused by TE111 mode in the DR. Fig. 4 (b) shows the simulated boresight gains with different

length of ls1. Two radiation nulls are generated both at the upper and lower band edges. The average gain decrease when the length of ls1 increase from 2.7 mm to 3.0 mm. A large deviation to the gain curve can be seen at the frequencies corresponding to the first and second resonance modes. The gain at the frequency corresponding to the third resonance mode keeps a relevant smaller change

Fig. 5 shows the simulated boresight gains and reflection coefficients with different length of ls2. The coupling between the two SIW cavities increases when the length of ls2 increases from 2.5 mm to 2.8 mm. It is also verified that the first two resonances are caused by the SIW cavities. Different from Fig. 4 (b), the average gain is not significantly affected with increase of ls2, as shown in Fig. 5 (b).

Based on the above discussion, the coupling between the SIW cavities and between DR and SIW cavity can be independently controlled by changing the length of ls1 and ls2. The first and second resonant mode is cause by SIW cavities. The last one is caused by the DR. Besides, two radiation nulls are generated both at the upper and lower band edges of the passband which could further improve the frequency selectivity of the proposed filtering DRA.

II. RESULTS AND DISCUSSION

Fig. 6 shows the simulated reflection coefficient and boresight gain of the proposed antenna. The impedance band width is 10.3 % from 24.7 GHz to 27.4 GHz with reflection coefficient less than -10dB which is corresponding with the 5G mm-Wave band at 28 GHz. An average gain of 6.7 dBi in the operating frequency band is achieved. Two radiation nulls are generated both at the upper and lower band edges. The simulated radiation patterns of the proposed filtering DRA at the frequencies of 24.7, 26.2 and 27.4 GHz are shown in Fig. 7, respectively. As seen in the figures, the proposed filtering DRA achieves symmetrical and stable radiation patterns and a cross-polarization lower than -30 dB in E-plane and -20 dB in H-plane.

CONCLUSION

In this letter, a differentially SIW TE20-mode fed substrate integrated filtering DRA has been presented. A filtering function was realized and two radiation nulls were induced at both upper and lower band edges. The proposed filtering DRA could be fabricated with standard PCB technology forming a substrate integrated antenna which would minimize the assembling errors. With differential feeding using field distribution of SIW TE20 mode, the proposed antenna has achieved a low cross-polarization and

(c)

Fig. 7. Simulated radiation patterns of the proposed filtering DRA. (a) 24.7 GHz. (b) 26.2 GHz. (c) 27.4 GHz.

symmetrical and stable radiation patterns. The design method presented in this letter is applicable to design a differential filtering DRA for mm-Wave applications.

REFERENCES

[1]M. J. Marcus, “5G and ‘IMT for 2020 and beyond’ [Spectrum policy and regulatory issues],” IEEE Wireless Commun., vol. 22, no. 4, pp. 2-3, August. 2015.

[2]R. D. Gupta and M. S. Parihar, “Differentially fed wideband rectangular DRA with high gain using short horn,” IEEE Antennas Wireless Propag. Lett., vol. 16, pp. 1804-1807, 2017.

[3]H. Tang, C. W Tong and J. X Chen, “Differential dual-polarized filtering dielectric resonator antenna,” IEEE Trans. Antennas Propag., vol. 66, no. 8, pp. 4298-4302, Aug. 2018.

[4]Y. M. Pan, K. W. Leung and K. M. Luk, “Design of the millimeterwave rectangular dielectric resonator antenna using a higher-order mode,” IEEE Trans. Antennas Propag., vol. 59, no. 8, pp. 2780-2788, Aug. 2011.

[5]C. K. Lin and S. J. Chung, “A compact filtering microstrip antenna with quasi-elliptic broadside antenna gain response”, IEEE Antennas Wireless Propag. Lett., vol. 10, pp.381-384, 2011.

[6]Y. Chen, W. Hong, Z. Kuai, and H. Wang, “Ku-band linearly polarized omnidirectional planar filtenna,” IEEE Antennas Wireless Propag. Lett., vol. 11, pp. 310–313, 2012.

[7]X. Y. Zhang, W. Duan, and Y.-M. Pan, “High-gain filtering patch antenna without extra circuit,” IEEE Trans. Antennas Propag., vol. 63, no. 12, pp. 5883–5888, Dec. 2015.

[8]H. Chu, J. Chen, S. Luo and Y. Guo, “A millimeter-Wave filtering monopulse antenna array based on substrate integrated waveguide technology,” IEEE Trans. Antennas Propag., vol. 64, no. 1, pp. 316321, Jan. 2016.