диафрагмированные волноводные фильтры / 0d567cb1-f4ab-4492-8042-cd0ca794b96c

Abstract—A two-pole half-mode substrate integrated waveguide (HMSIW) cavity-based filtering antenna is presented. The HMSIW cavity is used as a radiating resonator with a conventional substrate integrated waveguide (SIW) cavity functioning as a high-Q resonator of the filter, leading to its compact size. The concept of source-load coupling is introduced for filtering antennas, which allows for the presence of two controllable transmission zeroes (TZs) in the frequency response of proposed design. This is accomplished by using electric coupling between two resonator cavities and a combination of microstrip/ coaxial line to feed the antenna. The single layer printed circuit board (PCB) structure provides a broadside gain of 4.3 dBi at the operating frequency of 5.5 GHz. Gain curves are plotted at the broadside of the antenna, clearly showing a quasi-elliptic response and a 3-dB bandwidth of 2.6%.

Index Terms—Filtering Antenna, Substrate Integrated Waveguide (SIW), Half-mode Resonator, Quasi-elliptic Response.

I. INTRODUCTION

ILTERING antennas combine the radiating properties of Fantennas with frequency selective nature of microwave filters [1]. Co-design of filtering network with a radiating element as a constituent resonator not only leads to compact structures but also removes the transition losses that accompany the connection of a standard filter with an antenna. These dual-function devices usually employ a low-Q radiating element as the terminal resonator of the filtering network [1]– [4]. Furthermore, implementing filtering antennas in substrate integrated waveguide (SIW) technology provides high-Q selective nature of 3-D cavities in planar form [1]–[3]. Most of the SIW planar designs presented in the literature have multilayer integration of SIW cavities with radiating elements [1], [2]. In [3], a single layer Chebyeshev SIW filtering antenna with cavity backed slot antenna as the radiating element has been developed. Also, as described in [2], [3], the feeding microstrip lines also produce spurious radiation and affect the out-of-band characteristics of the filter. Metallic shields have been used in [2], [3] to remove the unintended effect of microstrip lines.

In this paper, a half-mode SIW (HMSIW) cavity resonator [5] is used in conjunction with a conventional SIW cavity to obtain a single-layer two-pole filtering antenna design. Electric coupling is introduced by using a coplanar waveguide (CPW) to couple the two cavities [6]. Source-load coupling

K.Dhwaj, J. M. Kovitz, H. Tian and T. Itoh are with the Department of Electrical and Computer Engineering, University of California, Los Angeles, CA, 90025 USA e-mail: kdhwaj@gmail.com

L.Jiang is with Department of Electrical and Electronics Engineering, University of Hong Kong, Pok Fu Lam, Hong Kong

Fig. 1. (a) Schematic of the proposed filtering antenna (top view) with a1 = 26 mm, a2 = 13mm, e1 = 26mm, e2 = 26mm, d1 = d2 = 1.6 mm, L1 = 8.5 mm, L2 = 2.8 mm, L3 = 4.5 mm, = 45 , t1 = 2 mm, t2 = 0.5 mm, t3 = 2.4 mm. (b) Side view with h = 0:79mm.

is introduced by using a radiating microstrip line to feed the antenna. By doing so, a transmission zero (TZ) is generated on either side of the passband leading to a quasi-elliptic frequency response. The position of TZs is then controlled by varying the feeding microstrip line length and terminating it by a coaxial line. The proposed HMSIW-integrated filtering antenna offers compact size and low cross-polarization, while maintaining reasonable antenna gain and efficiency values. The design methodology along with simulation and measured results of the proposed filtering antenna are presented in the following sections.

II. THEORY

Shown in Fig. 1 is the schematic of proposed filtering antenna. The antenna is designed for Rogers RT/Duroid 5880 board with r = 2.2 and tan = 0.001. The filtering antenna is composed of two resonators — a high Q conventional SIW cavity and a radiating HMSIW cavity — connected in an in-line topology through a coplanar waveguide (CPW) transmission line. The filtering antenna is fed by a 50 microstrip line, which in turn, is fed by a 50 coaxial line to minimize the reflection losses between the two.

Fig. 2. Lumped element model of the filter network. Parameters: C1 = 3.884 pF, C2 = 1.966 pF, L1 = 0.2149 nH, L2 = 0.425 nH, Gs = 0.02 S, GL = 0.01 S, JS;1 = 8.114 x 10 3 S, J2;L = 3.697 x 10 3 S, J1;2 = 2.22 x 10 3 S, JS;L = - 8.5 x 10 4 S.

Fig. 3. Frequency response of the circuit model.

Fig. 4. Variation of resistance Rr and normalized susceptance of the HMSIW cavity resonator.

Fig. 5. (a) HMSIW cavity resonator and (b) its transmission-line model with

YL = 1=Rr + jB.

The HMSIW cavity resonator is formed by removing half of the conventional SIW cavity through its symmetrical plane such that the T M110 field pattern in the remaining cavity is largely unaltered [5]. The open plane discontinuity, however, functions as a radiating discontinuity and is used in this design to provide terminating resistance to the two pole filtering network. Moreover, compared to a conventional SIW cavity, the HMSIW cavity provides a size shrinkage of more than 50%.

The lumped-element equivalent circuit for a two stage inline filter is shown in Fig. 2, where the LC tank circuits are used to model the two cavity resonators and admittance inverters are used to model the interstage couplings. The direct source-to-load coulping due to radiation from feeding network is modeled by the admittance inverter, JS;L. The lumped-element circuit model can be obtained by optimization methods shown in literature [7]. In this paper, the lumpedelement prototype is built for a 2.8% bandwidth filter with an in-band ripple level of 0.15 dB, operating frequency as 5.5 GHz and a TZ on either side of the passband (@5.25 GHz and @5.8 GHz) (Fig. 2). Response of the circuit model is shown in Fig. 3. Once the lumped-element values of the equivalent circuit are obtained, the constituent elements of the filtering antenna can be designed.

A. Conventional SIW Cavity

The resonance frequency of the conventional SIW resonator in the T M110 mode can be obtained as:

Fig. 6. Variation of (a) resonance frequency of the HMSIW cavity. (b) the admittance inverter JS;L

where c0 is the speed of light in air. The values of capacitances C1 and inductance L1 can be calculated as [7]:

B. HMSIW Cavity

In case of the HMSIW cavity resonator, the radiating edge presents a lumped capacitance C due to fringing fields and a lumped resistance, Rr to account for the far-field radiation.

Although Rr and C can be evaluated using analytical methods [8], an EM-solver (Ansys HFSS) has been used to de-embed the values for the proposed design. The variation of Rr and normalized susceptance of waveguide aperture B=Y0 (where B = 2 f C and Y0 is the characteristic admittance of the HMSIW) with the parameters h and e2 is shown in Fig. 4.

The resonance frequency of HMSIW cavity, f2 can be evaluated from the transmission-line model (Fig. 5) by using the condition ImfYing = 0, where

with as guided wave phase constant for the dominant mode of HMSIW, given by:

Fig. 6 shows the variation of resonance frequency of HMSIW cavity resonator. It can be seen that the calculated f2 values closely follow the simulated ones.

For calculating the values of L2 and C2, the HMSIW cavity is first terminated by a magnetic wall at its open end. The corresponding resonator lumped element values are given as:

and where f20 is the resonance frequency of HMSIW cavity terminated by magnetic wall and can be calculated using (3) with B = 0. As there is no radiation from the HMSIW in this condition, Rr = 1. Now, if on removing the magnetic wall, B=Y0 << 1, then the slight change in resonance frequency of HMSIW cavity resonator , f20 - f2 , can be attributed toC with the inductive component of tank resonator remaining constant, i.e.

with C = ((f20 f2)=f20) 2C20. Also, from Fig. 2 and Fig. 5, it can be deduced that

C. Interstage Couplings

The values of admittance inverter parameters can be related to external quality factor Qe and coupling coefficient k as

The values of JS;L can be controlled by changing the length of microstrip feed-line (Fig. 6 (b)).

The transmission poles are introduced in the frequency response of filtering antenna when circuit admittance (Y 0in) seen from the source in Fig. 2 meets the condition, Y 0in = GS

Freq (GHz)

Fig. 7. TZ variation with microstrip feed line length.

Fig. 8. Use of metallic shield to suppress the spurious radiation from microstrip feed line. Shown in the figure is only the shield and the top metal plane of the filtering antenna.

If the value of source-to-load coupling is small, its effect on the passband characteristics is not significant and as such, the pole locations can be controlled by changing the cavity resonator susceptances or the interstage couplings.

On the other hand, the TZs are generated when voltage on load, VL = 0. This condition requires the signal propagating through the cavity resonators to be cancelled at the load by that propagating directly from source-to-load through JS;L in Fig. 2. From an EM point of view, this means cancellation of farfield radiation from HMSIW edge in the broadside direction by that produced by the microstrip feed-line and the its inset in the SIW cavity, leading to broadside directivity, D = 0. The positive sign of interstage coupling in Fig. 2 refers to electric coupling, which is achieved by using a CPW line to couple the two cavities as shown in Fig. 1. As such, the positions of TZs can be altered by varying the microstrip feed-line length (Fig. 7) and terminating it with a non-radiating coaxial line. This requires the radiating microstrip line to be considered a part of antenna. It can also be noticed in Fig . 7 that the TZs vanish if a metallic shield is kept on top of the feeding microstrip line and its inset (Fig. 8). Presence of a shield prohibits radiation from microstrip line, thereby removing the source-to-load coupling.

III. MEASUREMENTS

The reflection coefficient of the filtering antenna is measured using an Agilent 8510C Vector Network Analyzer while the gain values are measured in a spherical near-field chamber at the University of Hong Kong. Simulated and measured responses of the filtering antenna compare closely with each other (Fig. 9). The antenna is centered at 5.5 GHz with a gain of 4.3 dBi in both simulation and measurement curves. On the

Fig. 9. Frequency response of the filtering antenna in broadside direction. Here, Gain = (1 jS11j2) x e x D.

Fig. 10. (a) E-plane and (b) H-plane radiation patterns of the filtering antenna at 5.5 GHz. Note that copol/xpol components follow Ludwig’s 3rd definition of cross-polarization

other hand, bandwidth of the measured structure is reduced to 2.6% from the intended 2.8% due to fabrication tolerances. The measured reflection coefficient is less than -20 dB at the operating frequency. Moreover, the filtering antenna shows the presence of two transmission zeroes (TZs), leading to a quasielliptic response.

Finally, the radiation patterns of the filtering antenna at the operating frequency are shown in Fig.10. It can be seen that the cross-polarized gain level remains 15 dB below the copolarized gain in both E- and H-planes. The simulated and measured directivities at the broadside are 5.15 dBi and 4.8 dBi respectively. These directivities along with the values of gains mentioned previously, lead to measured and simulated radiation efficiencies, e of 89% and 82% for the filtering antenna (including the microstrip feed-line). Fig. 11 shows the variation of measured efficiency with frequency. Finally, the proposed design is compared with other published filtering antennas in Table I, which brings out the twin advantages of controllable TZs and compact size of the proposed design.

IV. CONCLUSION

A two pole quasi-elliptic filtering antenna is proposed which is implemented by integrating HMSIW cavity resonator with a conventional SIW cavity. Simulated and measured results along with a circuit model for the filtering antenna are pre-

Fig. 11. Variation of radiation efficiency with frequency. Here, total efficiency, eT = (1 jS11j2) x e.

TABLE I

COMPARISON OF FILTERING ANTENNAS (* : BROADSIDE GAIN, v :

SIMULATED, S/L : SOURCE-LOAD COUPLING)

sented. TZ control is facilitated by using a combination of microstrip and coaxial line to feed the antenna. The singlelayer SIW filtering antenna offers small size and low crosspolarization in addition to reasonable gain and efficiency values.

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