диафрагмированные волноводные фильтры / babbebc2-9550-42e0-b55e-d385c1fdfd37

Figure 7 represents measured reflection coefficient S11 at the input ports of the transmission line shown in Figure 6. The experimental data confirm the shift in the resonance frequency of the transmission line toward the lower frequencies as the order of the CRCRs increases. Furthermore, from Figure 8 reprerepresenting measured transmission coefficient S21 of the transmission lines, the band-stop behavior of the transmission lines at around the resonance frequency of the CRCRs is observed. Similar to the simulation data in Figure 4, it is observed that by increasing the order of the CRCRs, the band-stop becomes narrower and shifts toward the lower frequency band.

Figure 8 represents measured phase of the transmission coefficient UðS21Þ of the transmission lines shown in Figure 6. The measured data confirm that the transmission lines loaded with CRCRs exhibit longer electrical length in the frequency regime below the resonance frequency of the CRCRs as the order of the CRCRs increases.

4. CONCLUSION

In summary, a comparative study on tuning the electrical length of the loaded transmission line by controlling the distributed capacitance of the rose-curve resonators through the resonators’ order was presented. Full-wave numerical simulations and laboratory experiments were carried out to validate the study.

ACKNOWLEDGMENT

The authors acknowledge the software support provided by CMC Microsystems.

REFERENCES

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14.O. Abu Safia, L. Talbi, and K. Hettak, A new type of transmission line-based metamaterial resonator and its implementation in original applications, IEEE Trans Magn 49 (2013), 968.

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16.A. Pradeep, S. Mridula, and P. Mohanan, Design of an edge-coupled dual-ring split-ring resonator, IEEE Antennas Propag Mag 53 (2011), 45–54.

17.S.I. Maslovski, P.M.T. Ikonen, I. Kolmakov, S.A. Tretyakov, and M. Kaunisto, Artificial magnetic materials based on the new magnetic particle: Metasolenoid, Prog Electromagn Res Pier 54 (2005), 61–81.

18.A. Kabiri, L. Yousefi, and O.M. Ramahi, On the fundamental limitations of artificial magnetic materials, IEEE Trans Antennas Propag 58 (2010), 2345–2353.

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VC 2017 Wiley Periodicals, Inc.

A COMPACT CPW-FED ANTENNA WITH FRACTAL S-SHAPED PATCHES FOR MULTIBAND APPLICATIONS

Hau Ran Cheong, Kim Ho Yeap, Koon Chun Lai, Peh Chiong Teh, and Humaira Nisar

Faculty of Engineering and Green Technology, Universiti Tunku Abdul Rahman, Jln Universiti, Bdr Barat, Kampar, Perak 31900, Malaysia; Corresponding author: laikc@utar.edu.my

Received 29 July 2016

ABSTRACT: We present the design of a novel compact coplanar waveguide-fed antenna with fractal s-shaped patches for multiband applications. The antenna consists of three fractal s-shaped patches with different lengths. We demonstrate that the number of resonant frequencies is in direct proportion with the number of fractal s-shaped patches. This is to say that, the number of resonant frequencies in a multiband antenna can be increased by increasing the number of patches. The resonant frequencies can be selected by carefully adjusting the length of the patches. We have designed our antenna to operate at 2.5/5.3/7.1/ 8.4 GHz. The measurement results show that the proposed antenna has 10 dB impedance bandwidths of 622 MHz (2.322–2.944 GHz), 466 MHz (5.113–5.579 GHz), 121 MHz (6.890–7.011 GHz) and 1080 MHz (8.206–9.286 GHz) to cover all the 2.4 GHz Bluetooth, 5.5 GHz WiMAX, and 2.4/5.2 GHz WLAN bands. The latter two bands of our proposed antenna fall within the X band range which finds vast applications in radar, aircraft, spacecraft and mobile or satellite communication system. The proposed antenna is printed on a single-layered FR4 substrate, and it occupies a small volume of 17 3 18 3 1.6 mm3. The simulated and

measured performance of the antenna confirms its omnidirectional radiation pattern and quad-band operation. VC 2017 Wiley Periodicals, Inc. Microwave Opt Technol Lett 59:541–546, 2017; View this article online at wileyonlinelibrary.com. DOI 10.1002/mop.30344

Key words: multiband; quad-band; monopole antenna; coplanar waveguide (CPW)-fed

1. INTRODUCTION

With the rapid growth in wireless communication systems, there has been significant improvement in the design of multiband antennas. Different designs for multiband antennas, applied particularly in Wireless Local Area Network (WLAN) and Worldwide Interoperability for Microwave Access (WiMAX), can be found in [1–14]. It can be seen from these literatures that most of the antennas are only designed to support dual-band [1,2], triband [3,7] or ultra wideband (UWB) [8–10] operation. Although the designs in [11–14] support up to four operating bands, the antenna structures are either complicated and, therefore, difficult to be fabricated or the designs take up unnecessary space, resulting in large sizes. Here, we present the design of a novel compact multiband patch antenna, which can support above three operating bands. The antenna is fed by a coplanar waveguide CPW transmission lines and the geometry of the antenna constitutes fractal s-shaped patches. Besides being compact and low profile, we will demonstrate that our s-shaped patch antenna exhibits good impedance and radiation characteristics when operating at 4 resonant frequencies, namely 2.5, 5.3, 7.1, and 8.4 GHz. Although a quad-band design would be illustrated in detail here, it is worthwhile noting that our antenna design has the flexibility of supporting above four operating bands. The number of bands and their respective resonant frequencies can be adjusted by varying the number and size of the s-shaped patches.

Figure 2 Geometry of the antenna with a single s-shaped patch. [Color figure can be viewed at wileyonlinelibrary.com]

2. ANTENNA DESIGN

Figure 1 presents the geometry of the proposed multiband monopole antenna fed by a CPW transmission line. The antenna is printed on an FR4-epoxy substrate of thickness 1.6 mm, loss tangent 0.022 and permittivity 4.4. The basic structure of the antenna consists of a radiating patch which is in fractal s- shaped, a ground plane, and a 50 X CPW feed line connected to a 50 X SMA connector. Figure 2 depicts the basic building block of the antenna which consists of only a single s-shaped patch. Ansoft high frequency structure simulator HFSS [15] has been used to determine the reflection coefficient of the structure. As can be seen from the simulated result in Figure 3, the single s-shaped patch antenna generates two resonant modes. Figure 4 depicts the structure of the antenna with a second s-shaped patch added to the original building block; whereas Figure 5 shows the corresponding reflection coefficient of this structure. It can be observed from the simulated result that the number of resonant modes has increased to four, i.e. twice of that obtained from a single s-shaped patch. The result indicates that, with every additional s-shaped patch appended to the original building block, the number of resonances will then be doubled.

The resonant frequencies fr depend on the lengths of the upper horizontal lines wi found on the s shapes, where the subscript i denotes the number of these lines. Because the fractal s- shaped patch in Figure 1 comprises 7 upper horizontal lines, wi ranges from w1 to w7. It is to be noted, however, that i can be of any number and is not only restricted to 7. It essentially depends on the number of resonant modes we wish to generate. By carefully varying the lengths of wi, the resonant frequencies

Figure 4 Geometry of the antenna with a two fractal s-shaped patches. [Color figure can be viewed at wileyonlinelibrary.com]

Figure 5 Simulated reflection coefficient of the antenna with two fractal s-shaped patches

fr can then be adjusted to the desired operating frequencies. Using HFSS, the optimum dimensions of the proposed antenna are determined. Figures 6 and 7 show respectively, the geometries of the proposed antenna, before and after optimization. As can be observed from Figure 8, the resonant frequencies for the six resonant modes fall exactly at 2.5/5.3/7.1/8.4 GHz after optimization. The final dimensions of the proposed antenna are specified in Table 1.

Figure 6 Geometry of the proposed antenna before optimization. [Color figure can be viewed at wileyonlinelibrary.com]

Figure 7 Geometry of the proposed antenna after optimization. [Color figure can be viewed at wileyonlinelibrary.com]

Figure 8 Simulated reflection coefficient of the proposed antenna, before (dashed line) and after optimization (dotted line). [Color figure can be viewed at wileyonlinelibrary.com]

3. RESULTS AND DISCUSSION

The design of our proposed antenna was fabricated, as shown in Figure 9. The measurement S11 parameters obtained from the

Figure 9 Photograph of the fabricated antenna. [Color figure can be viewed at wileyonlinelibrary.com]

Figure 10 Simulated (dotted line) and measured (solid line) reflection coefficient of the proposed antenna.

vector network analyzer VNA were compared with those found via simulation. Figure 10 illustrates the comparison between the simulation and measurement results. It can be seen that both results agree very closely with each other. It is worthwhile noting that, due to the close proximity between resonant frequency fr 58.4 GHz and 9.7 GHz, the bands for these two resonant modes are found to have combined in the measurement result, resulting in a wider bandwidth in that range. The measurement results show that the proposed antenna has 10 dB impedance bandwidths of 622 MHz (2.322–2.944 GHz), 466 MHz (5.113– 5.579 GHz), 121 MHz (6.890–7.011 GHz), and 1080 MHz (8.206–9.286 GHz). The former two resonant bands effectively cover all the 2.4 GHz Bluetooth, 5.5 GHz WiMAX, and 2.4/ 5.2 GHz WLAN bands; whereas the latter two bands fall within

TABLE 1 The Dimensions of the Proposed Antenna

the X band range which finds vast applications in radar, aircraft, spacecraft and mobile or satellite communication system [14].

To have a better insight of the return loss, we simulated the surface current density of the proposed antenna at 2.5, 5.3, 7.1, and 8.4 GHz. As shown in Figure 11, the highest surface current density concentrates at different parts of the structure at different resonant frequencies fr. At fr 52.5 GHz, the highest current density is found to have concentrated on lines w2, w3, w4, and w5. On the other hand, the highest surface current density con-

trate on two lines, i.e. w3 and w4 and at 8.4 GHz, it concentrates on lines w3 and w7. Close inspection on the figure, it could also be observed that the surface current at fr 55.3 GHz has the highest magnitude and it distributes widely throughout the antenna structure. The strong surface current suggests strongly that fr 55.3 GHz obtains the highest wave coupling among all

four resonant frequencies and it, therefore, shows the lowest reflection coefficient in Figure 10.

Figure 12 shows the simulated radiation patterns of the proposed antenna at 2.5, 5.3 7.1, and 8.4 GHz along the E- and H- planes. It can be observed from the figure that the radiation patterns are almost omnidirectional at the four resonant frequencies. It can be seen from Figure 13 that the obtained gains are

Figure 13 Simulated realized gain of fractal S-shape antenna

about 211.32, 24.91, 22.51, and 25.38 dB in the desired bands.

4. CONCLUSION

In this article, a compact novel CPW-fed antenna with fractal s- shaped patch for multiband applications has been proposed and successfully fabricated. The multiband characteristic is achieved by introducing the fractal s-shape as the radiating stub. The experimental results show that the antenna produces 10 dB impedance bandwidths at four bands—from 2.322 to 2.944 GHz, from 5.113 to 5.579 GHz, from 6.890 to 7.011 GHz and from 8.206 to 9.286 GHz. The antenna consists of three fractal s-shaped patches with different lengths. By increasing the number of fractal s-shaped patches, the number of resonant bands can be increased proportionately. Also, by adjusting the horizontal lengths of the s-shaped patches, the resonant frequencies can be adjusted to the desired frequencies. The proposed antenna is easy to fabricate and has a simple configuration. It also gives good impedance and radiation characteristics. Hence,

the proposed antenna can be a good candidate for multiband applications.

REFERENCES

1.J.H. Yoon and G.S. Kil, Compact monopole antenna with two strips and a rectangular-slit ground plane for dual-band WLAN/WiMAX applications, Microwave Opt Technol Lett 54 (2012), 1559–1566.

2.A. Moradhesari, M.N. Moghadasi, and F.G. Gharakhili, Design of compact CPW-fed monopole antenna for WLAN/WiMAX applications using a pair of F-shaped slits on the patch, Microwave Opt Technol Lett 55 (2013), 2337–2340.

3.J. Pei, A.G. Wang, S. Gao, and W. Leng, Miniaturized triple-band antenna with a defected ground plane for WLAN/WiMAX applications, IEEE Antennas Wireless Propag Lett 10 (2011), 298–301.

4.H. Chen, X. Yang, Y.Z. Yin, S.T. Fan, and J.J. Wu, Triband planar monopole antenna with compact radiator for WLAN/WiMAX applications, IEEE Antennas Wireless Propag Lett 12 (2013), 1140–1143.

5.L. Dang, Z.Y. Lei, Y.J. Xie, G.L. Ning, and J. Fan, A compact microstrip slot triple-band antenna for WLAN/WiMAX applications, IEEE Antennas Wireless Propag Lett 9 (2010), 1178–1181.

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8.M. Ojaroudi, M. Hassanpour, C. Ghobadi, and J. Nourinia, A novel planar inverted-F antenna (PIFA) for WLAN/WiMAX applications, Microwave Opt Technol Lett 53 (2011), 649–652.

9.A.M. Asghar, M. Malick, M. Karlsson, and A. Hussain, A multiwideband planar monopole antenna for 4G devices, Microwave Opt Technol Lett 55 (2013), 589–593.

10.A. Valizade, C. Ghobadi, J. Nourinia, and M. Ojaroudi, A novel design of reconfigurable slot antenna with switchable band notch and multi-resonance functions for UWB applications, IEEE Antennas Wireless Propag Lett 11 (2012), 1166–1169.

11.A. Dadgarpour, A. Abbosh, and F. Jolani, Planar multiband antenna for compact mobile transceivers, IEEE Antennas Wireless Propag Lett 10 (2011), 651–654.

12.J. Pourahmadazar, C. Ghobadi, J. Nourinia, and H. Shirzad, Multiband ring fractal monopole antenna for mobile devices, IEEE Antennas Wireless Propag Lett 9 (2010), 863–866.

13.H. Lavakhamseh, C. Ghobadi, J. Nourinia, and M. Ojaroudi, Multiresonance printed monopole antenna for DCS/WLAN/WiMAX applications, Microwave Opt Technol Lett 54 (2012), 297–300.

14.M.A. Motin, M.I. Hasan, M.S. Habib, and M.I. Sheikh, Design of a modified rectangular patch antenna for quad band application. In Informatics, Electronics & Vision (ICIEV), 2013 International Conference on (pp. 1–4). IEEE, Vancouver 2013, May.

15.Ansoft Corporation, Ansoft High Frequency Structure Simulation (HFSS), Ver. 13, Ansoft Corporation, Pittsburgh, PA, 2010.

VC 2017 Wiley Periodicals, Inc.

2.4-GHz CMOS LINEAR POWER AMPLIFIER FOR IEEE 802.11N WLAN APPLICATIONS

Jinho Yoo, Changhyun Lee, Inseong Kang, Minoh Son, Yonghoon Sim, and Changkun Park

Soongsil University, 369 Sangdo-ro, Dongjak-gu, Seoul 06978, Republic of Korea; Corresponding author: pck77@ssu.ac.kr

Received 9 August 2016

ABSTRACT: In this work, we design a linear CMOS power amplifier with a spiral-type output transformer for IEEE 802.11n WLAN applications. We conduct studies to identify the proper output transformer and

power stage structures for linear CMOS power amplifiers. The power amplifier is composed of a single differential-pair for the power stage to mitigate the stability problems that frequently arise in high gain linear power amplifiers. Additionally, we investigate the output matching network using a spiral-type output transformer to minimize the output return loss. To verify the feasibility of the power amplifier, we designed a 2.4-GHz power amplifier with a 180-nm SOI CMOS process. The designed power amplifier is measured using an IEEE 802.11n WLAN signal. The power amplifier achieves 21.28 dBm output power while the measured EVM satisfies the standard for 802.11n applications. VC 2017 Wiley Periodicals, Inc. Microwave Opt Technol Lett 59:546–550, 2017; View this article online at wileyonlinelibrary.com. DOI 10.1002/ mop.30343

Key words: cascode; CMOS; differential; amplifier; transformer

1. INTRODUCTION

Demand for fully-integrated transceivers including a RF power amplifier for WLAN solutions has recently increased in efforts to realize low-cost system solutions [1–3]. Accordingly, fullyintegrated RF front-ends including RF power amplifiers for WLAN applications are considered an essential technology. To realize fully-integrated RF front-ends, CMOS based power amplifiers are necessary. In tandem with recent advances in CMOS technology, many advances in the development of CMOS power amplifiers have been introduced [4–7]. In particular, nonlinear CMOS power amplifiers using voltage combining methods have been actively studied [8–10]. A distributed active transformer (DAT) is regarded as one of the most effective techniques to realize the voltage combining method [8]. By virtue of the DAT, transformers have become one of the most widely used elements of output matching networks in CMOS power amplifiers. Furthermore, various meaningful studies related to the transformer structure for improved efficiency have been reported [11–14]. However, although use of the DAT is one of the effective techniques for the nonlinear CMOS power amplifier to obtain high efficiency, further investigation of the transformer structure associated with stability problems in linear CMOS power amplifier is required.

In this work, we investigate the proper output transformer and power stages for linear CMOS power amplifiers to mitigate the stability problems associated with high gain linear CMOS power amplifiers. Additionally, we investigate the output matching network using a spiral-type transformer to minimize the output return loss.

Figure 1 Simple block-diagram of the typical CMOS power amplifier using DAT